Wide-Angle Beam Scanning Phased Array Antennas: A Review

Phased array antennas (PAAs) with flexible and agile wide-angle beam scanning (WABS) abilities have drawn rapidly increasing attentions in recent years. They have found significant applications in beyond fifth-generation (B5G) and sixth-generation (6G) wireless communications, satellite communications, sensing, etc., for their outstanding capability to achieve wide spatial coverage, enable critical frequency reuse, and improve the system capacity. In this review, we aim to investigate and discuss several challenges that hinder WABS for conventional PAAs, including the strong mutual coupling, narrow beamwidth of the element antenna, etc. We then review and summarize a variety of innovative techniques to overcome these challenges. Subsequently, we discuss and analyze potential research gaps of WABS PAAs for future emerging applications. This timely review fulfills the demand of a thorough and detailed investigation on WABS technologies for PAAs and provides detailed introduction and guidance for interested researchers and engineers.

SECTION I. Introduction

Phased array antennas (PAAs), comprising of two or more antenna elements fed coherently with controllable phase shifters to achieve desired pattern performance such as beam scanning and shaping, is a crucial technology for modern communications and sensing systems. The PAA behaves like a choir for a music concert. Each member in the choir, i.e., the antenna element in the PAA, should be well coordinated to produce an insistent high-performance rhythm. PAAs have numerous advantages, and they are capable of achieving high gain, low sidelobe level (SLL), flexible beam scanning, and high tracking accuracy.

Particularly, PAAs capable of agile beam scanning within a wide angular range is a critical enabler for current and future terrestrial and non-terrestrial wireless communications networks [1], [2]. Wide-angle beam scanning (WABS) PAAs can find many significant applications; some of them are shown in Fig. 1. Modern wireless communications have become an ubiquitous part of our daily lives and activities, which results in ever-increasing data traffic and device connections [3]. The data traffic in wireless networks is expected to grow even faster. The beyond fifth-generation (B5G) and sixth-generation (6G) communications are promised to meet the demands in the foreseeable future. They can realize data rates at a level of gigabit per second (Gbps), and bring millisecond-level of latency time, far higher traffic volume density, super dense connections, and improved efficiencies in terms of energy and cost [4]. WABS antennas can deliver critical frequency reuse and significantly improved system capacity, which has been considered as a key enabler technology for B5G and 6G communications [5], [6], [7], [8].

FIGURE 1.

Several application scenarios of wide-angle beam scanning (WABS) phased array antennas (PAAs).

Over the past decade, we have also witnessed an increasing demand for satellite communications (SATCOM) [9]. Some commercial companies like Starlink, OneWeb, and Kuiper have already begun their race of launching space-based broadband communication satellites. In SATCOM, electronic beam steering PAAs are highly desired since they generally have lower profiles than lens or reflector antennas and do not occupy large space as the mechanical scanned arrays. Moreover, with the development of the silicon technology in the past decade, silicon beamformer chips have been employed in large-scale PAAs for SATCOM to reduce the cost and to offer a high degree of integration [9], [10], [11]. Thus, highly integrated PAAs with WABS ability that can achieve global coverage are becoming very promising in SATCOM, especially for medium earth orbit (MEO) and low earth orbit (LEO) SATCOM.

Another important application of WABS PAAs is the sensing-based automotive radar and anti-collision radar. Automotive radar is like the eye of the automobiles; it is a key technology to achieve autonomous driving. It can make driving safer and more convenient by relieving the drivers from monotonic tasks and split-second decisions within complex traffic conditions. WABS PAAs are highly promising candidates given their high-gain scanning beams in a very wide angular range [12], [13]. Furthermore, WABS PAAs can also find applications in airborne platforms like unmanned aerial vehicles (UAVs) [14] and aircrafts [15], synthetic aperture radar [16], etc. This wide range of applications has largely driven the evolution and development of WABS PAA technologies.

The focus of this work is to review and summarize main challenges and recent advances for facilitating WABS with PAAs. Although some literatures have been reported to achieve this objective, there are still some limitations and potential gaps. In [17], Kavitha and Raglend briefly introduced the wide-angle scanning PAAs and reviewed several types of antenna elements suitable for WABS. But the challenges to achieve WABS were not thoroughly discussed, and only a few technologies were discussed. In [18], Li et al. briefly discussed the main challenges of achieving WABS and presented a review of wideband WABS. However, they focused on only wideband WABS PAAs with target applications of SATCOM. Besides, it is worth mentioning that a special issue on the low-cost WABS antennas was published recently in [19]. In this special issue, a variety of technologies were developed and reported to achieve low-cost technologies in the theory, design, development, measurements, and in-field deployment of the WABS PAAs. Based on the above state-of-the-art, we present a comprehensive and timely review on WABS PAAs in this paper. We will mainly discuss the challenges and technologies to achieve WABS for linear and planar PAAs. Conformal PAAs, such as cylindrical PAAs that can achieve very wide scanning thanks to their curved structures, are out of the scope of this review. The main contributions of this work are summarized as follows. Firstly, we have comprehensively discussed the main problems and challenges encountered in achieving WABS for linear and planar arrays. In addition, we thoroughly reviewed innovative techniques and strategies to tackle these challenges by category according to their main contributions. Moreover, future research potential gaps are adequately discussed in this paper. It is expected that this paper will facilitate the research, development and technology uptake of WABS arrays in current and future communications and sensing systems.

This paper is organized as follows. In Section II, the background of PAAs is briefly discussed. Section III presents the main challenges in facilitating WABS. In Sections IV–​VII, the state-of-the-art achieving WABS by utilizing different technologies are thoroughly reviewed. Section VIII describes the future research potential gaps. Section IX draws the conclusion.

SECTION II. Brief Introduction to Phased Array Antennas

In this section, the PAA is briefly introduced. Then some popular optimization strategies for designing PAAs to achieve desired radiation performances are summarized.

A. Phased Array Antennas

PAAs are those that can achieve desired beam pattern performance by individually controlling phase shifters connected to each array element with or without excitation amplitude modulation. PAA scanning is a modern beam scanning technology that evolves from conventional mechanical scanning arrays. It is known that the mechanical scanning arrays have to mechanically rotate antenna arrays to steer their main beams. As a consequence, they suffer from many severe issues that hinders their applications for modern wireless systems, such as having a low scanning speed, being bulky and lack of accuracy, etc. On the contrary, benefiting from the use of controllable phase shifters, PAAs usually have a fast scanning speed and high accuracy, and exhibit a high reliability and flexibility in beam steering and shaping.

Fig. 2 shows a schematic view of a typical PAA with a planar array arrangement. As noted, this planar PAA has a total of $M \times N$ individual elements distributed in the xoy-plane. Each element in the array is connected with a phase shifter that provides the requisite phase for beam steering. With the array setup as shown in Fig. 2, the far-field pattern $F (\theta, \phi)$ of this PAA can be expressed as: $$\begin{equation*} F \left ({\theta, \phi }\right) = \sum _{m=1}^{M} \sum _{n=1}^{N} I_{m,n} E_{m,n}\left ({\theta,\phi }\right) e^{j\left \{{ k_{0} \vec {r}_{m,n} \cdot \vec {u}\left ({\theta,\phi }\right) + \alpha _{m,n}}\right \}}\tag{1}\end{equation*}$$ where $j=\sqrt {-1}$ ; $k_{0} = 2\pi /\lambda$ is the wavenumber in free space; $\theta$ is the angle measured from positive $z$ -axis and $\phi$ is the azimuth angle measured from positive $x$ -axis; $E_{m,n}(\theta,\phi)$ denotes the pattern of the element in the $m$ th $(m = 1, 2,\ldots, M)$ row and $n$ th $(n = 1, 2,\ldots, N)$ column; $\vec {u}(\theta,\phi) = \sin \theta \cos \phi \vec {e}_{x}+\sin \theta \sin \phi \vec {e}_{y} + \cos \theta \vec {e}_{z}$ is the propagation direction vector. $\vec {r}_{m,n}$ , $I_{m,n}$ , and $\alpha _{m,n}$ are the location, excitation amplitude, and excitation phase of the $(m,n)$ th element, respectively. It is common knowledge that one generally can control the excitation phases $\alpha _{m,n}$ and amplitudes $I_{m,n}$ to achieve desired pattern performance of PAAs.

FIGURE 2.

Schematic view of a typical planar phased array antenna (PAA) with phase shifters for beam scanning.

B. Optimization Strategies

Generally, besides the intuitive beam scanning, PAAs should be equipped with additional performance optimization for practical applications, such as having low sidelobe levels (SLLs), achieving low cross-polarization levels, suppressing grating lobes, etc. To accomplish the above goals along with the beam scanning, one can optimize the array parameters associated with the PAAs, including the excitation amplitudes and phases, and element positions.

Optimizing both excitation amplitude and phase is a common approach as reported in the literature. Numerous excitation amplitude and phase optimization methods have been developed in the past decades, including the iterative Fourier transform (IFT) and its variants [20], [21], [22], convex optimization techniques [23], [24], alternating projection methods [25], [26], stochastic algorithms [27], [28], and machine learning methods [29], [30], [31]. They have achieved scanned beams with excellent performance in terms of mainlobe shape, SLL reduction, and null notching in specified directions.

The phase-only optimization method is another effective approach for PAA pattern synthesis. It avoids using nonuniform excitation amplitudes, thus leading to simplified feeding networks since amplitude weighting requires unequal power dividers. A lot of effective methods were developed in the past to design PAAs employing phase-only optimization [32], [33], [34], [35]. For example, in [34], an adaptive bat algorithm was developed to achieve pattern nulling at directions of interferences by optimizing the excitation phases only. It was shown that nulls can be precisely imposed to an arbitrary interference direction using the developed bat algorithm-based beamformer.

Over this decade, element rotation was included as a brand new degree of freedom to replace amplitude weighting in the PAA synthesis [36], [37], [38], [39], [40], [41], [42], [43], [44]. In comparison to conventional amplitude weighting, the method of using element rotation can greatly simplify the feeding network and the associated cost and complexity for the PAAs by saving many unequal power dividers [42]. With the element rotation, beam scanning patterns can be achieved with reduced SLLs [37], [38]. Besides, axial ratio for the circular polarization pattern can be improved with the element rotation technique [39], [40]. Moreover, shaped beam patterns with scanning can also be realized with the developed methods in [42], [43], [44].

SECTION III. Challenges for Wide-Angle Beam Scanning (WABS)

Achieving WABS for PAAs has been one of the most challenging tasks. Specifically, good impedance matching and consistent gains should be maintained when the beam is scanned in a wide angular range. The wide angular range of PAAs is usually interpreted as being larger than 100°. In the realization of WABS PAAs, there are three key issues to be considered as follows.

The first is the beamwidth of antenna element pattern. Theoretically, gain pattern of an array is the product of the array factor and the element pattern when each element is assumed to have identical radiation pattern [45]. Therefore, for an antenna element with a traditional broadside radiation pattern, the array peak gain will decrease as the beam scans away from broadside due to the gain drop of the element pattern [46]. As an example, a 12-element linear array with uniform inter-element spacing of half wavelength is considered. Its element pattern is assumed to be $\cos \theta$ . Fig. 3 shows the array patterns being normalized to the broadside gain when the scanning angles are 0°, +20°, +40°, and +60°, respectively. As noted, the peak gain of one specifically beam-steered array pattern is proportional to the element gain at its main beam angle. As a consequence, the array gain will drop dramatically at large scanning angles if the element pattern has a narrow half-power beamwidth (HPBW).

FIGURE 3.

Beam scanning patterns for a 12-element linear array with the element pattern of $\cos \theta$ .

In order to mitigate large gain drops at large scanning angles, the element pattern is expected to have a broad HPBW. This can be verified through a beam scanning array reported in [47]. Its element antenna has HPBWs of 93° and 113° in the E-plane and H-plane, respectively. The array configuration with this element exhibited a 4.7 dB gain decrease with respect to the broadside gain when scanning to +45° in the E-plane. Whilst the gain decreases only 2.3 dB when the beam is scanned to +45° in the H-plane.

The second issue is the mutual coupling among PAA elements. It is known that array elements are usually spaced at a relatively small distance (around $0.5\lambda$ ) to avoid appearance of the grating lobes when the main beam is scanned. With a small element space, the radiation characteristics of one element in the array would be generally affected by the electromagnetic environment consisting of all the other elements. This phenomenon is called mutual coupling effect. The mutual coupling will not only affect the beam shape of the element pattern, but also cause its impedance matching to deteriorate. Moreover, the impedance characteristics would be varied with a dependence on the beam scanning angles. As a result, it would be a formidable task to match the impedance in a wide scanning angle range. Scan blindness is a common issue that occurs for WABS PAAs [48].

The third factor that should be carefully handled is grating lobes. Although the grating lobe can be avoided by setting a relatively small element spacing, a large element spacing is inevitable in some particular applications. These include the size of the element antenna being larger than half a wavelength, the mutual coupling effect being severe when the spacing is smaller than half a wavelength, etc. With a large spacing, the grating lobe may appear in the visible region especially when the array beam is scanned to large angles. Thus, suppressing grating lobe is of great importance to increase array gain and efficiency [45]. Theoretically, to avoid grating lobes in a uniformly linear or rectangular-distributed planar array, the element spacing $d$ should meet $$\begin{equation*} d \leq \frac {\lambda }{1+ \left |{\sin \theta _{max}}\right |}\tag{2}\end{equation*}$$ where $\theta _{max}$ is the maximum scanning angle. If $\theta _{max}$ is anticipated to be 60°, $d$ should be maintained smaller than $0.53\lambda$ . Otherwise, grating lobes will appear in the visible region. Special techniques should thus be introduced to suppress grating lobes when a larger spacing is used.

The above challenging issues have attracted great research efforts along with significant advances for WABS PAAs. In the following sections, we will review and discuss the latest developments.

SECTION IV. Wide-Beamwidth Element Antenna

A number of innovative studies have been reported to maximize the beam scanning range of PAAs by developing and employing broad-beam antenna elements. There are two main antenna types being reported to facilitate the element pattern with wide beamwidth. The first type is passive single-port element antennas, and their beamwidths are intentionally broadened by sacrificing directivity and gain. This is comprehensible given the fact that the beamwidth is generally inversely proportional to the gain/directivity of the antenna. The other type is beam switchable element antennas. These include pattern reconfigurable element antennas and multi-port element antennas. Using this technique, the overall coverage of those available beams will become very large even though the beamwidth of each pattern is not wide. However, the switching will increase the cost and complexity, and there exists time delay for beam switching. In this section, we will review and discuss these two types of wide-beamwidth antenna elements.

A. Wide-Beamwidth Passive Single-Port Element Antenna

We will mainly discuss two popular techniques to realize the wide-beamwidth passive single-port element. These include image theory-based element antenna and multimode element antenna. In addition, we will also briefly introduce some other methods for achieving passive wide beamwidth.

1) Image Theory Based-Element Antenna

Generally speaking, an electromagnetic image antenna consists of an electric source positioned above a large perfect electric conducting (PEC) ground plane, or a magnetic source positioned above a large perfect magnetic conducting (PMC) ground plane. Owing to the inherent wide beamwidth nature of these electric/magnetic sources as well as the introduced proper imaging effect, the antenna beamwidth will be very broad. This technique is quite popular and has been widely reported in [13], [49], [50], [51], [52], [53], [54], [55], [56]. In the image-based antenna, the radiator can be equivalent to electric current $\boldsymbol {J}$ or magnetic current $\boldsymbol {M}$ ; the ground plane has the PEC type and PMC type; the relative orientations of the radiator with reference to the ground plane can be parallel or perpendicular [49]. By using different combinations of the radiators and the ground planes, one can obtain eight basic antenna types. These eight types of antennas and their theoretical patterns are shown in Fig. 4. Take the $\boldsymbol {M}$ pE-antenna as an example, the “ $\boldsymbol {M}$ ” indicates the radiator is an equivalent magnetic current; “p” means the relative position is parallel; “E” indicates that the ground plane is PEC. Among the eight types of antennas, the radiation patterns $F(\theta)$ of the $\boldsymbol {M}$ pE- and $\boldsymbol {J}$ pM-antennas can be expressed as [49]: $$\begin{equation*} F(\theta) = \cos \left ({k_{0} H \cos \theta }\right)\tag{3}\end{equation*}$$ where $H$ is the distance between the equivalent current source and the ground plane. As inferred from (3), the radiation patterns of $\boldsymbol {M}$ pE- and $\boldsymbol {J}$ pM- antennas are omnidirectional in the upper-half space as shown in Fig. 4(c) and (d). They are ideal candidates for serving as the element of WABS arrays.

FIGURE 4.

Eight basic types of image based antennas with (a) a PEC ground and (b) a PMC ground, and the corresponding theoretical radiation patterns (c) and (d) [49].

An artificial magnetic conductor (AMC) backed antenna was designed as a $\boldsymbol {J}$ pM-type antenna in [49] to achieve WABS. As shown in Fig. 5(a), the antenna is equivalent to a dipole above an AMC ground. The HPBW of this antenna element in the H-plane is about 160°, covering almost the whole upper half space. Subsequently, an 11-element array was constructed, and it achieved a wide angle beam scan from −89° to +90° in the H-plane, as shown in Fig. 5. In [50], besides WABS, the $\boldsymbol {J}$ pM AMC-backed dipole antenna was further improved to reduce the radar cross Section (RCS) of the antenna array.

FIGURE 5.

Geometry of the AMC-backed dipole antenna (a), and (b) the scanning patterns of the constructed 11-element linear array at 5.8 GHz in [49].

The $\boldsymbol {M}$ pE-type antenna can be constructed by simply placing a slot antenna on a large PEC ground plane (also known as magnetic dipole antenna) [13], [52], [53], [54], [55], [56]. For example, Liu et al. developed a microstrip magnetic dipole antenna with an omnidirectional radiation pattern in the upper half-space for WABS in [55]. The antenna configuration is shown in Fig. 6(a). An equivalent magnetic current source was created through the gap between the two pads, and an $\boldsymbol {M}$ pE-type antenna was obtained. Then, an E-plane and an H-plane array were respectively constructed with this $\boldsymbol {M}$ pE-type element. For the E-plane array, the main beam was scanned from −76° to +76° as shown in Fig. 6(b). On the other hand, a beam scanning range from −64° to +64° was achieved for the H-plane array. As later reported by the same research group, the $\boldsymbol {M}$ pE-type element was miniaturized by developing an optimized wire antenna with a significantly reduced width [56]. The modified antenna achieved equivalent magnetic current performance. Subsequently, a 9-element linear array was constructed using the wire antenna, which enabled a beam scanning range from −75° to +75°.

FIGURE 6.

Structure of the developed magnetic dipole antenna (a), and (b) the measured WABS radiation patterns for the constructed 9-element E-plane array in [55].

2) Multimode Element Antenna

Rather than using a single mode for radiation, the multimode element antenna can excite and combine several resonant modes to realize a broadened beam [57], [58], [59], [60], [61], [62], [63]. A wide-beam microstrip antenna with metal walls that works at 3.2 to 3.8 GHz was reported in [57]. It consists of a U-slot patch antenna and vertical electric walls over the ground plane. A horizontal current is produced on the radiating patch, and a vertical current is induced on the electric walls driven by E-fields of the patch antenna, as can be seen in Fig. 7(a). A broadened beamwidth is achieved by combining the horizontal and vertical currents, as shown in Fig. 7(b). A 9-element H-plane scanning linear array and a 9-element E-plane scanning linear array were then constructed, which achieved a beam scanning range from −90° to +90° in the H-plane and from −75° to +75° in the E-plane. Other designs that achieve wide-beamwidth by inducing vertical currents using vertical electric walls or metallic strips were also reported in [58], [59], [60].

FIGURE 7.

The antenna with vertical walls reported in [57] and its current distribution as well as radiation patterns. (a) shows side view of the antenna with the two types of currents. (b) shows the radiation patterns of this antenna.

It is noticed that the method of introducing vertical currents will inevitably increase the profile of an antenna. To maintain a low profile, multimode patch antennas would be more preferred [61], [62], [63]. A single-fed hybrid-mode patch antenna was developed in [61] that achieved a wide beamwidth of 128°. As shown in Fig. 8(a), a TM₁₀ mode and TM₂₀ mode can be excited at the left sub-patch and the whole patch simultaneously. When these two modes are combined, a wide-beam pattern is realized. An 8-element linear array was built with this antenna element, and a beam scanning range from −68° to +66° was achieved with gain fluctuation smaller than 1.5 dB, as shown in Fig. 8(b).

FIGURE 8.

Geometry of the dual-mode antenna (a) reported in [61], and (b) the measured beam scanning patterns of the constructed $1 \times 8$ -element linear array.

Besides the above developments, dual-polarization behavior that is necessary for some applications was incorporated into the wide-bandwidth antenna element in some reported works. A dual-polarized wide-beamwidth antenna element was reported in [62]. It consists of a circular patch and four rectangular parasitic patches distributed around the circular patch, as shown in Fig. 9(a). Dual polarization was realized by switching the feeding positions $P_{1}$ and $P_{2}$ . A TM₁₁ mode was excited in the circular patch, and a zero-order resonant (ZOR) mode was excited in the parasitic mushroom-like rectangular patches. Ideally, an omnidirectional pattern will be obtained when these two modes are combined. This was validated by the simulated and measured patterns of this antenna as shown in Fig. 9(b). The developed wide-beam element was used to form an $8\times 8$ planar PAA with 2-D WABS performance in [62], and a WABS range of about ±65° in the two orthogonal planes with gain fluctuation less than ±1.73 dB was achieved. The xoz-plane beam scanning patterns are shown in Fig. 9(c). Another dual-polarized antenna consisting of a mushroom-shaped patch and two circular patches employing the ZOR and TM₀₁₀ modes was developed in [63]. The combination of the two modes achieved a radiation pattern with a wide coverage of up to 125°. A $6\times 6$ -element planar PAA was designed using the developed antenna. Dual-polarized beam scanning was realized in a WABS range of ±66° in both xoz and yoz planes with a scan loss of 3.5 dB.

FIGURE 9.

Top view of the antenna (a) reported in [62], and (b) its radiation patterns as well as (c) the simulated beam scanning patterns of the constructed $8 \times 8$ -element planar array at xoz-plane.

3) Other Methods

In addition to the aforementioned two popular methods, there are also some other techniques to accomplish wide-beamwidth passive antenna elements [64], [65], [66], [67], [68], [69], [70], [71], [72]. For example, a parasitic pixel layer-based antenna was reported in [65]. Fig. 10(a), (b), and (c) show the antenna configuration. It has two copper layers: a lower layer with an E-shaped patch and a ground plane; an upper layer with $4\times 4$ small patches arranged periodically to pixelate the metal surface on the substrate. Strong coupling between the driven antenna and the parasitic surface induced large currents on the surface. In the parasitic surface, the connection between these small patches was appropriately selected for wide-beam performance by using a co-simulation strategy with full-wave simulation and multi-objective optimization. It is shown that the HPBWs of the antenna are 188° and 156° in xoz- and yoz-planes. An $8\times 8$ -element planar array was then constructed, and it achieved a beam scanning range of ±75° in xoz-plane, as shown in Fig. 10(d).

FIGURE 10.

Structure of the reported antenna in [65] and the array beam scanning patterns. (a) Lower layer. (b) Upper layer. (c) Side view. (d) Beam scanning patterns of an $8 \times 8$ -element array in xoz-plane.

Magnetic-electric dipole (ME-dipole) and Huygens sources are also effective in broadening the beamwidth of an antenna. For example, a circular polarized ME-dipole in [66] and a metamaterial-loaded DRA that integrates a pair of out-of-phase Huygens sources in [67] were presented for wide-angle beam scanning. Beam scanning ranges of ±66° and ±65° were achieved, respectively. A DRA with a saddle-shaped radiation pattern was developed in [68]. Its element pattern has a gain that is nearly inversely proportional to the peak gain of the array factor across a wide angular range and, consequently, the total array gain can stay almost identical in the whole scanning range. A 9-element linear array based on this DRA was designed, and a ±72° beam scanning was obtained. A hybrid topology optimization method was reported in [69] to optimize the shape of an patch antenna to achieve wide radiation beamwidth. For a 9-element linear array constructed with this element, a scanning range of ±80° was achieved.

Furthermore, high impedance surfaces (HISs) were deployed to realize the wide beamwidth in a low-profile fashion, as reported in [70], [71], [72]. A high-impedance periodic structure (HIPS) is proposed as a replacement for the traditional PEC ground of a U-slot loaded patch antenna in [71]. The HIPS results in an improvement in surface wave behavior, which in turn enhances the beamwidth of the antenna element pattern. The proposed antenna was used to form a 9-element linear array, which achieved a wide angle range of ±60° with only a 3.3 dB gain fluctuation. Mushroom-like HIS was used in [72] as the ground of a dipole antenna to broaden the element pattern beamwidth. The final array exhibits a wide scan angle up to ±85°.

B. Beam Switchable Element Antenna

Pattern reconfiguration is one commonly used method to actively achieve wide beamwidth for element antenna. A number of relevant works employing this technique have been reported [14], [73], [74], [75], [76], [77], [78], [79], [80], [81], [82], [83], [84], [85]. For example, a mode reconfigurable antenna employing PIN diodes working at 5.8 GHz was reported in [73]. It consists of a rectangular patch with 12 symmetrical slots for miniaturization and a reconfigurable feeding network, as shown in Fig. 11(a). By controlling the PIN diodes, the antenna can be switched between two different modes: TM₀₁ mode with a broadside beam and TM₂₀ mode with a conical beam, as respectively shown in Fig. 11(b) and (c). A $1\times 4$ phased array was constructed with this element, and the array achieved a beam scan from −75° to +75° in the elevation plane with a gain fluctuation less than 3 dB, as shown in Fig. 11(c) and (d).

FIGURE 11.

The mode configurable antenna reported in [73] with PIN diodes to switch between TM₀₁ and TM₂₀ modes (a), element patterns for the two modes (b) and (c), and measured beam scanning array patterns using the two modes (d) and (e) for a $1 \times 4$ -element array at 5.8 GHz.

One can also use antennas with multiple individual feeding ports to excite different modes with complimentary patterns for wide beamwidth [14], [74], [75]. An overall broad-beam pattern with dual orthogonal polarizations was achieved in [74] by combining three working modes: TM₁₀ and TM₀₁ modes by exciting a square patch and a TM₂₁ mode by exciting the circular ring patch. An $8\times 8$ planar array comprising of this element achieved a wide beam scanning range of ±64° in two orthogonal planes, and the beam scanning patterns in xoz-plane are shown in Fig. 12. In [75], a windmill-shaped loop antenna backed with an HIS was developed. The antenna employed an upper and a lower substrate, and a mushroom-like HIS comprised of rectangular patches and shorted vias was placed at the back of the lower substrate to eliminate the back radiation with reduced aperture. The antenna realized independently directed tilted beams by using four different excitation ports. A $4\times 4$ -element array was constructed with this antenna, and it was capable of scanning its main beam up to +72° in elevation in each of the four spatial quadrants.

FIGURE 12.

The measured beam scanning patterns in xoz-plane for a $8 \times 8$ -element array reported in [74]. (a) Scanning performance using TM₁₀ mode. (b) Scanning performance using TM₂₁ mode.

Note that the above multi-port antennas individually excite each port for a special beam pattern. By switching the excitation port, different beam patterns can be chosen and, hence, an overall wide beamwidth is obtained by combining all these operating states. An alternative method that manipulates the phases for multiple excitation ports can also achieve wide beamwidth [76], [77]. A dual-port pattern-reconfigurable antenna was reported in [76], as shown in Fig. 13(a), that uses reconfigurable phase shifters for beam reconfiguration. The pattern reconfiguration was achieved by exciting the two ports with phase differences of −90° and 90°, respectively. The final four-element PAA achieved a scanning range of ±162° with a gain fluctuation of 1.25 dB, as shown in Fig. 13(b). In addition, other investigations reveal that wide-angle beam scanning can be realized in several cutting planes [78], [80], [81]. A dielectric resonator antenna with eight reconfigurable horizontally tilted beams was developed in [78]. A ±60° beam scanning in multiple planes of $\phi = 0^{\circ }$ , 45°, 90°, and 135° was achieved by a $4\times 4$ -element array of the designed antenna.

FIGURE 13.

Configuration of the pattern reconfigurable DRA (a), and (b) the beam scanning radiation patterns of the $1\times 4$ linear array reported in [76].

C. Brief Summary

Based on the above discussions on the wide beamwidth element antenna technologies, the following findings can be summarized when comparing beam switchable antenna elements and passive broad-beam single-port ones in the context of WABS. Passive broad-beam elements exhibit the benefits of a simple feeding network and easy implementation. However, they generally have lower gains due to the trade-off made to achieve wider beamwidth. In contrast, beam switchable elements can provide higher gains but require additional control components or techniques for beam switching, such as PIN diodes. Therefore, the selection of antenna element can be determined based on custom-designed requirements for achieving WABS PAAs.

SECTION V. Mutual Coupling in Wide Angle PAA

Mutual coupling has a large effect on array beam scanning performance, especially at large angles. It is a crucial issue that should be carefully handled for WABS PAA [86]. Generally, there are two main methods to deal with mutual coupling issues: The first is considering, optimizing, and utilizing mutual coupling in the element design and the resultant PAA with this element can be naturally suitable for WABS. The second is to mitigate the mutual coupling among antenna elements and, hence, the mutual coupling effect can be negligible during WABS. In the following, we will discuss these methods and the associated antennas reported in the literature in detail.

A. Mutual Coupling Based Optimization

When mutual coupling exists in the PAA, the coupling coefficients generally vary when the array main beam scans to different angles. This is because the changing rate of active input susceptance at a high beam-steering angle is generally faster than that of the admittance during the beam scanning [87]. Therefore, implementing a wide-angle impedance matching (WAIM) layer to compensate for this difference is an effective method that can optimize the impedance matching in a very large angular range.

Many designs using pure dielectric or metasurface superstrates have been developed, such as those in [87], [88], [89], to realize WAIM. For example, a unit cell with a perforated dielectric sheet working as a WAIM layer was reported in [87]. The unit cell consists of a Vivldi-like antenna and a perforated WAIM, as shown in Fig. 14(a). A $16\times 16$ -element array was formed with this unit cell. When the beam was scanned to 70° in the E-plane and H-plane, the array showed an active S-parameter characteristic of less than −9.13 dB. The gain decreased by 3.83 and 2.87 dB for the E-plane and H-plane scanning, respectively. Fig. 14(b) shows the unit cell employing a metasurface WAIM layer with split rings reported in [88]. It has been demonstrated that, comparing to dielectric WAIM or the case without WAIM layer, this metasurface WAIM can increase the scan range in the D- and H-planes by 16° and 10°, respectively, while having a negligible impact in the E-plane.

FIGURE 14.

Structures of the unit cells with dielectric WAIM layer reported in [87] (a), and (b) the metasurface WAIM layer with split rings reported in [88].

Despite their outstanding performance in WABS, the above antenna arrays cannot achieve a very wide operating bandwidth. Further investigations were carried out to improve the bandwidth of WABS PAA. Tightly coupled antenna arrays (TCAAs) technology exploits the strong coupling capacitance between the array elements to counteract the inductive effect of the ground on the antenna. This way a wide bandwidth even ultra-wideband (UWB) operation can be obtained.

Electrical dipole is a popular unit structure being employed in TCAAs [90], [91], [92], [93], [94], [95], [96], [97], [98], [99], [100], [101], [102], [103], [104], [105], [106], [107]. For example, in [96], a balanced antipodal dipole was designed for UWB and WABS tightly coupled dipole array (TCDA). The unit configuration is shown in Fig. 15(a). As noted, a metasurface WAIM is placed above the dipole to achieve a wider angle scanning characteristic while maintaining a low profile. A $16 \times 16$ -element planar array was formed with this unit. Fig. 15(b) shows the attained VSWRs of the center element when the array has its beam scanned to different angles. Simulated and measured results verified that the developed TCDA was able to operate over a 5:1 bandwidth with active VSWR < {2, 2.8, 3.2} for scanning to {broadside, 45°, 60°}, respectively. In [97], a millimeter-wave (mmWave) TCDA operating in a wide bandwidth from 23.5 to 29.5 GHz was realized. It achieved a beam scanning up to ±60° in E-, H-, and D-plane within the whole operating band.

FIGURE 15.

Structure of the tightly coupled dipole array (TCDA) unit cell with a metasurface WAIM layer reported in [96] (a), and (b) the VSWRs of an center element in a $16 \times 16$ -element planar array for different scanning angles.

B. Mutual Coupling Mitigation

Rather than optimizing the mutual coupling between array elements, one can mitigate the mutual coupling and, hence, alleviate the mutual coupling effect for WABS PAAs. Several interesting methods have been reported to achieve this objective.

Electromagnetic bandgap (EBG) structures are generally 3-D periodic objects that prevent the propagation of electromagnetic waves in a specified band of frequency for all angles and for all polarizations [108]. Thus, EBG structures are effective in reducing mutual coupling and improving the WAIM since propagation of the surface waves in the bandgap of EBG can be suppressed due to high surface impedance [109]. A wideband wide-beam scanning PAA with cavity-backed antennas was developed in [110]. The detailed antenna structure with the EBG is shown in Fig. 16(a) and (b). The element was fed by a microstrip line through a feed patch with a backed cavity. Four narrow strips on a suspended thin substrate were placed at the top surface to obtain the best impedance matching over the entire frequency band. Mushroom-like EBG structures were utilized to suppress the E-plane scan blindness caused by the surface waves. A $13\times 5$ triangular distributed planar array was exampled. Good impedance matching was achieved at different scanning angles within the operating band of $7-13$ GHz. The active VSWRs of a center element for different scanning angles are shown in Fig. 16(c), and the beam scanning patterns in E-plane at 10 GHz is provided in Fig. 16(d).

FIGURE 16.

Geometries of the reported unit with EBGs and the corresponding element and array performances [110]. (a) 3-D view of the unit, (b) 3-D view of the EBG unit, (c) measured and simulated active VSWRs of a central element, and (d) the E-plane scanning patterns of a $13 \times 5$ -element triangular-distributed array at 10 GHZ.

Defected ground structure (DGS) is another commonly used strategy for achieving WAIM by reducing the mutual coupling [111]. By employing the DGSs, a metasurface-based antenna was reported in [112] to achieve WABS. Fig. 17(a) shows the slot-fed grid square patch metasurface antenna array. The DGS was realized by the meander-line slots etched on the ground plane. Benefiting from the DGS, the scanning range was much improved without increasing any dimension. The array can scan its main beam to a maximum angle of 50° at the 5.2 GHz, as shown in Fig. 17(b). Moreover, enhanced scanning capabilities can be obtained over the wide frequency band from 4.6 GHz to 5.8 GHz. Besides, in [113], a metasurface antenna with DGSs for scanning blindness suppression and impedance matching improvement was reported. It realized an $8\times 8$ metasurface PAA with ±60° beam scanning range and impedance bandwidth of 22.2%.

FIGURE 17.

Configuration of the developed $1 \times 9$ -element metasurface antenna array with DGSs (a), and (b) the corresponding scanning patterns in [112].

Apart from the above effective methods for mutual coupling mitigation, a concept of array-antenna decoupling surface (ADS) was developed in [114]. The ADS is a thin layer of low-loss and low dielectric constant substrate printed with a plurality of electrical small metal reflection patches. These reflection patches are carefully designed to create diffracted waves at the coupled antenna port to cancel the coupled waves meanwhile minimizing the perturbation to the original array antenna. The distance between the ADS and the ground plane of the array antenna is determined to make sure the diffracted wave is out of phase with respect to the coupled waves from the coupled antenna element port. Design guidelines and considerations of the ADS were discussed in [114], and two practical design examples were provided to evaluate the ADS. Fig. 18(a) shows the configuration of one exampled PAA. The representative S-parameters of antenna elements in the array with and without the ADS are given in Fig. 18(b) and (c), respectively. It is seen that the mutual coupling between any two adjacent elements, i.e., $S_{12}$ or $S_{23}$ , is significantly reduced from about −15 dB to below −30 dB, which has demonstrated the good performance of the ADS in reducing the mutual coupling.

FIGURE 18.

Eight-element patch antenna array with a designed ADS for mutual coupling reduction (a) and the representative measured S-parameters of the array with and without the ADS of (b) antennas 1 and 2 and (c) antennas 2 and 3 [114].

In addition, some other technologies such as the metasurface slabs [115], reactive impedance surfaces [116], and frequency selective surfaces (FSSs) [117] have been reported to suppress the mutual coupling for WABS.

C. Brief Summary

In relation to the above review, it is noted that the mutual coupling based optimization method is widely employed to facilitate WABS PAAs. This method can effectively incorporate and leverage mutual coupling for achieving WABS PAA. Furthermore, additional performance features such as wide operating bandwidth can be generated by optimizing the mutual coupling, as demonstrated by TCAAs. However, this optimization may result in a higher profile due to the introduction of WAIM layers. Regarding the mutual coupling mitigation method, employing an EBG or a DGS structure can maintain a lower profile for the PAA. However, when employing other mutual coupling mitigation techniques involving ADS, metasurface slab, reactive impedance surface, and FSS, the profile still remains an issue. It is clear that the method chosen for the WABS PAA would differ depending on whether the objective is to obtain a wide bandwidth or maintain a low profile.

SECTION VI. Grating Lobe Cancellation

In most practical applications, grating lobes should be avoided since they generally cause unwanted radiations, thereby degrading the gain and efficiency of the PAA [118]. As previously analyzed, grating lobes would appear especially when the beam is scanned to large angles. It is common knowledge that an array pattern is the product of the element pattern and array factor. Therefore, to eliminate the grating lobes of WABS PAA, two main methods have been widely implemented: one is employing nonuniform or triangular-latticed element distributions in the aspect of the array factor; the other revolves around the element pattern aspect and entails the use of antenna elements with pattern nulls precisely positioned at the grating lobe directions.

A. Grating Lobe Cancellation Methods

Using triangular-latticed array layout is a simple but effective way to avoid grating lobes. Compared to rectangular-latticed array, the triangular-latticed one has an enhanced spacing range that is free from the grating lobes apart from the advantage of element number reduction. To avoid grating lobes, the element spacing in an triangular-latticed array should meet [45] $$\begin{align*} d_{x}\leq&\frac {1}{\sin \alpha } \cdot \frac {\lambda }{1+ \left |{\sin \theta _{max}}\right |} \tag{4}\\ d_{y}\leq&\frac {1}{\cos \alpha } \cdot \frac {\lambda }{1+ \left |{\sin \theta _{max}}\right |} \tag{5}\end{align*}$$ where $$\begin{equation*} \pi /6 \leq \alpha \leq \pi /3 \tag{6}\end{equation*}$$ is the basic angle of the isosceles triangle in the triangular grid, $d_{x}$ and $d_{y}$ are element spacing in two dimensions, and $\theta _{max}$ is the maximum scanning angle. As indicated from (4) and (5), assuming $\theta _{max}=60^{\circ }$ is anticipated, the element spacing $d_{x}$ and $d_{y}$ should be maintained smaller than $0.76\lambda$ when $\alpha = 45^{\circ }$ . The value of $0.76\lambda$ is $0.23\lambda$ larger than that for the rectangular layout. Many interesting PAAs have been presented to achieve WABS by employing triangular-latticed layouts [87], [110], [116], [119], [120], [121]. Some examples of them are shown in Fig. 19. They can generally achieve beam scanning ranges no less than 60° with gain degradation smaller than 3 dB.

FIGURE 19.

WABS Arrays with triangular latticed elements for avoiding the grating lobes. (a) a $5\times 9$ -element array presented in [116], (b) a $2\times 8$ -element array-element array presented in [119], and (c) a $12\times 20$ -element array presented in [121].

Employing a nonuniform element distribution is also an effective method to alleviate the grating lobes. Such method can break the periodicity and redistribute the energy in the grating lobes throughout the far field pattern [122]. Many effective methods have been proposed to mitigate the grating lobes by employing nonuniform or sparse element spacing [123], [124], [125], [126]. For example, in [123], the genetic algorithm is employed to achieve thinned aperiodic linear PAAs to suppress the grating lobes with increased steering angles. The optimized array can achieve a beam scanning angle of 60° with constrained SLLs.

Another popular method for grating lobe cancellation is employing antenna elements with nulls or low power at the grating lobe directions [127], [128], [129], [130]. In [127], [128], [129], dual-mode antenna elements were designed to achieve element patterns with main beams at the anticipated scanning region and low power at the grating lobe directions. For example, in [129], a dual-mode antenna consisting of a square DRA with $TE_{1\delta 1}$ mode and a monopole with $TM_{101}$ mode was designed. These dual modes were excited with two individual ports. The radiation patterns of the dual modes are shown in Fig. 20(a). By exciting the dual modes simultaneously with identical amplitudes but a phase difference of $\delta _{01} = 180^{\circ }$ , as shown in Fig. 20(b), the overall element pattern with a peak in the scanning angle meanwhile a low power in the grating lobe direction was achieved. With this dual-mode antenna, a $1 \times 4$ -element linear array with an inter-element spacing of $0.95 \lambda$ was constructed. Fig. 20(c) and (d) show the beam scanning patterns. As is seen, grating lobe is well canceled by adopting the antenna element.

FIGURE 20.

Radiation patterns of a dual-mode element and the beam scanning patterns of a $1 \times 4$ -element $0.95 \lambda$ -spaced linear array in [129]. (a) radiation pattern of the dual mode (TE and TM modes) antenna. (b) the element patterns when its two modes are fed with equal amplitudes but different phase differences. (c) and (d) show the array pattern with scanning angles of 15°, 30°, 45°, and 60° with grating lobe cancellation. Patterns with the elements all working at TE mode are also shown for comparison.

In addition, a null scanning antenna with varactor loaded impedance reconfigurable circuits was developed in [130]. The exploded view of the designed antenna is shown in Fig. 21(a). The antenna can provide continuous null scanning for grating lobe cancellation as well as desired main beam steering in two orthogonal planes. The beam scanning of the antenna element pattern along with the array pattern for a $4 \times 4$ -element $0.8 \lambda$ -spaced planar array are shown in Fig. 21(b). Array factor without element pattern is also provided for comparison. It is shown that, by adopting this antenna element, the grating lobe can be effectively suppressed during the scanning by keeping the null always at the grating lobe direction.

FIGURE 21.

The null scanning antenna as well as the element and array patterns presented in [130]. (a) exploded view of the null scanning antenna, and (b) the element pattern and array patterns of a $4 \times 4$ -element $0.8 \lambda$ -spaced planar array with different scanning angles. (b) also shows the array factor with isotropic element pattern as a comparison.

B. Brief Summary

In summary, achieving grating lobe cancellation can be approached from the perspective of either the element pattern or the array factor. From the array factor perspective, both nonuniformly-spaced and triangular-latticed layouts are effective methods. The former method may require a more complicated feeding network while the latter one can simplify the feeding network. However, the grating lobe cancellation effect is comparatively limited with the triangular-latticed layout. Regarding the element pattern perspective, using beam switching antenna elements with null patterns can effectively suppress the grating lobe. However, the introduction of PIN diodes or varactors can lead to additional losses and reduced efficiency. The associated biasing network will also be complex. Interested readers should carefully evaluate these factors and select appropriate techniques for grating lobe cancellation in WABS PAAs.

SECTION VII. Frequency Scanning PAA for Wide-Angle Beam Scanning (WABS)

We have thoroughly discussed the electronically controlled PAAs for WABS in the previous sections. As reported, the electronically scanned PAAs showed attractive radiation and impedance performance with great flexibility. However, those performance generally need to use many phase shifters to controlling the excitation phases of all the elements. As a consequence, the complexity as well as the system cost is greatly increased. To this end, frequency-based beam scanning is a promising choice for low-cost beam scanning systems. They do not rely on any phase shifters for beam scanning. In contrast, they achieve beam scanning by changing the source frequency of the PAA. It is evident that the frequency scanning PAA has many significant advantages such as being simple, easy for fabrication, and low-cost. This makes the frequency scanning PAA being attractive for many modern applications including imaging systems and direction finding networks [131], [132]. It is also acknowledged that owing to the dependence on the operating frequency, the freedom for beam scanning is much reduced in comparison to that with phase shifters. This may lead to performance degradation for frequency scanning PAAs compared to the electronically controlled PAAs. Moreover, the variation of frequency at different beam angles prohibits the PAA application in many systems such as most communications systems. Therefore, there is a trade-off between the system cost and the performance characteristics.

Fig. 22 shows a systematic view of the leaky-wave antenna (LWA), one typical type of the frequency scanning PAA, for WABS. The LWA employs guiding structures to support wave propagation, and the electromagnetic energy is gradually radiated or leaked with the wave traveling along the waveguiding structures [133]. The guiding wave is represented by the long connected line and the radiators can be treated as an array of individual antenna elements, as seen in Fig. 22. Thus, the LWA is equivalent to a series feed antenna array. For one-dimensional (1-D) LWA, the main beam angle $\theta$ can be obtained through the following relation [134], [135]: $$\begin{equation*} \theta (f) = \sin ^{-1} \frac {\beta (f)}{k_{0}(f)} \tag{7}\end{equation*}$$ where $\beta$ is phase constant in the waveguiding structure. Generally, the traveling wave has an inherent dispersion, which contributes to a frequency-based beam scanning for 1-D LWA. For example, an 1-D slot array LWA was reported in [136] as shown in Fig. 23(a). It has both longitudinal and transverse slots and is loaded with an array of shorting vias in each unit cell. By sweeping the operating frequency band $(|S_{11}| \leq -10$ dB) from 8.5 to 14.4 GHz, the antenna achieved a measured beam scanning range from −53° to +37°, as observed in Fig. 23(b) and (c).

FIGURE 22.

A schematic view of the leaky-wave antenna for WABS.

FIGURE 23.

1-D slot array leaky-wave antenna reported in [136]. (a) The fabricated antenna prototype. (b) Simulated and measured S-parameters. (c) Measured beam scanning radiation patterns.

To facilitate WABS with LWAs, there are several critical challenges. The first is an open-stop band (OSB) issue. The OSB is caused by the reflection from each unit cell adding up in-phase at the input port when the beam points towards the broadside direction [137]. It leads to deteriorated impedance matching and degraded gain. Therefore, the OSB effect prevents a continuous beam scanning through broadside for LWAs. To suppress the OSB, researchers have developed numerous techniques, including utilizing composite right/left handed (CRLH) structures [136], [138], [139], [140], [141] and employing impedance matching methods [142], [143], [144]. The second are those difficulties we have introduced and discussed for the electronically scanned PAA in the previous sections. For example, one needs to realize wide element pattern beamwidth to compensate for gain losses at large angles and maintain wide-angle impedance matching to avoid large gain variations.

To date, a variety of techniques have been reported to overcome the above challenges and, hence, to facilitate LWAs with frequency-based WABS. A slotted SIW LWA with partially reflecting vias was reported in [131], and its configuration is shown in Fig. 24. Unlike the conventional SIW with two closely-spaced via walls (acting as PEC walls), the developed antenna has one via wall sparsely distributed for OSB suppression. The specified-shaped slot was used and optimized to improve the linear polarization purity. Fig. 24 shows the simulated and measured S-parameters. It is seen that the OSB was mitigated and a wide operating bandwidth was obtained. By sweeping the frequency from 7.625 to 11 GHz, the main beam was scanned from −74° to +45°, a total range of 119°. As revealed in [132], the beam scanning range can be further increased, being from −76° to +67°. This was accomplished by employing a combination structure of composite right/left-handed (CRLH) lines, magnetic dipole, and alumina ceramic block. Note that these WABS LWAs can’t point towards endfire directions. One main reason is the effect caused by the ground plane. In this regard, those ground-free structures are promising to further enhance the beam scanning range from backward to forward and further to endfire directions. Various techniques including the Goubau line [145], [146] and spoof surface plasmon polariton (SSPP) [147], [148], [149] were reported to achieve that goal. In [148], double-layer glide-symmetric SSPP structures were developed that scans the beam from −90° to +44.3°. The attained realized gain ranges from 8.03 to 12.17 dBi. Besides the elevational beam scanning as reported by the above WABS LWAs, the periodic co-planar-strip LWA reported in [150] realized an azimuthal beam scanning range from −75° to +84°. The azimuthal directed beam was obtained by using open-ended slots.

FIGURE 24.

Slotted SIW LWA with partially reflecting vias and its performances reported in [131]. (a) The fabricated antenna prototype. (b) Simulated and measured S-parameters. (c) Measured beam scanning radiation patterns.

SECTION VIII. Discussion and Future Scope

In Section III, we discussed the main challenges in facilitating WABS. In Sections IV–​VI, we conducted a comprehensive review of the existing literature, presenting various techniques to address these challenges. Notably, we did not distinguish between linear and planar arrays in those discussions. This is because both types of arrays face similar challenges, and the techniques employed to tackle these challenges are also comparable. For example, both linear and planar arrays need to be cautious of the appearance of grating lobes as the element spacing increases. They also need to address mutual coupling issues, particularly at large scanning angles, and require antenna elements with relatively wide beamwidths for WABS. A slight difference is that mutual coupling in planar arrays is more severe than that in linear arrays. Furthermore, it is clear that planar arrays require the elements with wide beamwidths in dual or multiple cuts to achieve WABS in corresponding planes [65], whereas linear arrays only require wide beamwidths in one cut within the scanning plane.

Although WABS PAAs have achieved significant developments, there are several potential research challenges for future emerging systems with the evolution of wireless technologies. As discussed in Section IV, wide-beamwidth antenna elements are commonly utilized for WABS with small gain decrease at large angles. However, this wide-beamwidth element generally has a narrow bandwidth and, hence, the associated PAAs are also narrow-band [151]. Further efforts are needed to increase their bandwidths to broaden their areas of application. In addition, regarding the method of using beam switchable element, complex control and additional losses are unavoidable due to the involvement of switching devices/ports, as previously mentioned. Therefore, future research should aim to minimize the losses and complexities. Potentially, the element can be optimized to locate the switching devices in small-current areas while maintaining beam switching performance.

Tightly coupled arrays are generally used to achieve wideband or ultra-wideband WABS. These arrays typically require wideband feeding structures, like baluns, and impedance matching layers. This may lead to bulky configurations and high profiles. Further work is expected to optimize the configuration into an integrated and compact structure.

The future B5G and 6G communications demands antenna arrays that operate at mmWave and even higher frequencies to support high data rates and to achieve low latency. However, compared to their lower frequency counterparts, there has been limited research on mmWave and even higher-band WABS PAAs. Higher frequencies pose new challenges due to the small physical sizes of antenna elements, which result in difficulties with manufacturing feasibility, assembly complexity, and beamforming network complexity [152]. Additionally, losses can be very high at mmWave frequencies. Therefore, future research on mmWave WABS PAAs should prioritize improving the efficiency, reducing costs, and addressing fabrication feasibility.

Multibeam antennas are regarded as a critical technology for B5G and 6G wireless communications networks. Achieving wide-angle scanning multibeams is an important research focus for the future. Analog beamforming networks, which offer advantages such as low cost and low energy consumption, have gained significant interest in recent years. A recently proposed solution, the generalized joined coupler (GJC) matrix, allows for independent scanning of multibeams [153], [154]. As such, GJC matrices hold great promise for achieving wide-angle scanning multibeams. However, research on GJC matrices is still in its early stage, and further research is necessary to enable continuous wide-angle scanning multibeams.

SECTION IX. Conclusion

This paper presented a thorough review of wide-angle beam scanning (WABS) phased array antennas (PAAs) in the literature. It is shown that there are three crucial factors that should be considered for a WABS PAA: the beamwidth of element pattern, mutual coupling among elements, and grating lobes. To tackle these challenges, a number of developed technologies with electronically scanned PAAs have been discussed in this review, including implementing a wide beamwidth element antenna with passive or active methods, optimizing the mutual coupling with wide-angle impedance matching (WAIM) or mitigating the mutual coupling, and suppressing grating lobes. Finally, WABS frequency scanned PAAs featuring easy for fabrication and low cost were also investigated. This timely review fulfills the need for a comprehensive investigation of WABS technologies for PAAs, providing a thorough and detailed introduction and guidance for interested researchers and engineers. It is expected that the paper will stimulate further research and development on this important technology.