<![CDATA[ IEEE Transactions on Antennas and Propagation - new TOC ]]>
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TOC Alert for Publication# 8 2018April 26<![CDATA[Table of contents]]>664C11634381<![CDATA[IEEE Transactions on Antennas and Propagation]]>664C2C266<![CDATA[Norton Equivalent Circuit for Pulsed Photoconductive Antennas–Part I: Theoretical Model]]>664163516451615<![CDATA[Norton Equivalent Circuit for Pulsed Photoconductive Antennas—Part II: Experimental Validation]]>664164616594355<![CDATA[Dual-Band Leaky-Wave Antenna Based on Dual- Mode Composite Microstrip Line for Microwave and Millimeter-Wave Applications]]>10-like mode, can propagate along the transmission line structure simultaneously. The DMC-MSL with a dual-layered planar configuration combines the merits of the MSL and the SIW. In addition, by adjusting the permittivity of the dielectric laminate used for realizing the thin SIW, the frequency ratio of the two operating modes can be varied flexibly. Based on the DMC-MSL structure, a novel dual-band leaky-wave antenna is presented. Periodic parasitic microstrip patches and the transverse slot array etched on the SIW are introduced in order to generate the required spatial harmonics. A feed structure that can excite the two operating modes of the DMC-MSL separately is designed as well. The fabricated prototype verifies that backward and forward frequency-scanning radiation beams are achieved within the low-frequency band from 5.75 to 8.5 GHz and the high-frequency band from 35 to 41.5 GHz, respectively. The proposed DMS-MSL structure and the leaky-wave antenna offer a new mean to implement antennas and devices for microwave and millimeter-wave dual-band applications.]]>664166016682675<![CDATA[A Five-Port Integrated UWB and Narrowband Antennas System Design for CR Applications]]>$times36$ mm $times1.6$ mm. The UWB antenna coupled with port 1 is designed to sense the complete UWB spectrum of 3.1–10.6 GHz approved by Federal Communications Commission in 2002. The four NB antennas operate at different single or dual bands to cover the complete UWB spectrum. In particular, the first NB antenna allied with port 2 achieves a single operating band ranges from 8.7 to 9.92 GHz. The second NB antenna associated with port 3 also covers a single band from 9.82 to 10.74 GHz. The third NB antenna linked to port 4 yields dual bands from 3.06 to 4.23 GHz and 6.33 to 8.83 GHz. The fourth NB antenna accessed at port 5 operates in a single band from 4 to 6.37 GHz. Isolation between the antennas is less than 16 dB over the antennas operating bands. Finally, the proposed integrated structure is fabricated and verified. Experimental results have a good agreement with the simulated results.]]>664166916765616<![CDATA[Tapered Fed Compact UWB MIMO-Diversity Antenna With Dual Band-Notched Characteristics]]>$times$ 34 mm. The tapered microstrip fed slot Antenna acts as a single radiating element with inverted L-shaped slits to introduce notches at wireless local area network and the IEEE INSAT/Super-Extended C-bands. The mutual coupling of less than −22 dB is achieved over the entire operating band (2.93–20 GHz). At the center of notched band, the efficiency of the Antenna drops that indicates a good interference suppression performance. The performance of the MIMO Antenna in terms of isolation among the ports, radiation pattern, efficiency, realized gain, envelope correlation coefficient, mean effective gain, and total active reflection coefficient is studied.]]>664167716842438<![CDATA[A Wideband Beam-Steerable Slot Antenna Using Artificial Magnetic Conductors With Simple Structure]]>$lambda $ to achieve a wideband performance. By turning on/off the p-i-n diodes on these two parasitic slots, the main lobe of the radiation pattern can be steered. The AMC surface is used to make the antenna radiate directionally reducing the back lobe simultaneously. A design procedure of tuning the proposed antenna to other frequency band is also given. The simulation and measurement results, including $S_{11}$ , radiation patterns, and realized gains are illustrated. Results show that the main lobe direction of the radiation pattern of the proposed antenna can be tuned among 0°, −36°, and 36° with the peak gains of more than 7 dBi in the operating bandwidth from 4.9 to 5.5 GHz.]]>664168516943611<![CDATA[A Conformal Differentially Fed Antenna for Ingestible Capsule System]]>3 by notching meandering slots to achieve miniaturization. The simulated impedance bandwidth ($vert S_{11}vert < -10$ dB) covers from 861 to 942 MHz. The integrated capsule system, inserted in a cubic homogeneous muscle phantom for initial parametric studies and optimization, occupies the dimensions of 22 mm (length) $times11$ mm (diameter). Meanwhile, the performance on the sensitivity to surrounding environment and specific absorption rate distribution are studied. In the measurements, the conformal antenna is rolled around a 3-D-printed capsule inside minced pork. The pork temperature data are monitored in time and transferred wirelessly. To confirm the system reliability, hardware circuits are designed with differential concept and a data logger is also realized to record in-time temperature data. Finally, the feasibility of system function demonstrates the suitability of the conformal differentially fed antenna in biomedical applications.]]>664169517035908<![CDATA[Sidelobe-Reduced and Gain-Enhanced Square Patch Antennas With Adjustable Beamwidth Under TM<sub>03</sub> Mode Operation]]>03 mode is proposed in this paper to realize gain enhancement, sidelobe reduction, and adjustable beamwidth by reshaping the surface current distribution on the patch resonator. On the one hand, by introducing transverse slots along the central line of patch, the medial part of out-of-phase longitudinal current is interrupted to circulate with little radiation. In this way, the patch serves as a full-wave dipole rather than a $3lambda $ /2 one, and thus, the sidelobe level (SLL) in the E-plane is substantially reduced. On the other hand, the slots are installed in proximity to the nonradiative edges and at the patch center, respectively, to arouse distinctive effect on the current densities across the width of patch. Consequently, the beamwidth in the H-plane can be freely adjusted by different configurations of these slots. At last, a square patch antenna with three cascaded slots loaded along the central line is designed, fabricated, and tested. The simulated and measured results show that the directivity of a single proposed patch antenna can achieve as high as 13.3 dB with the SLL reduced to about −20 dB.]]>664170417133439<![CDATA[A New Approach to Microwave Rectennas Using Tightly Coupled Antennas]]>$4times 4$ TCA array was simulated, fabricated, and tested. Numerical simulations showed that the finite array has the potential to achieve a radiation-to-ac power conversion efficiency of 90% and a radiation-to-dc power conversion efficiency of 72%, while laboratory measurements gave a maximum radiation-to-dc power conversion efficiency of approximately 60%.]]>664171417243921<![CDATA[Fragmented Antenna Realization Using Coupled Small Radiating Elements]]>$12 times 10times 10$ cm ($0.096lambda _{0}times 0.08lambda _{0}times 0.08lambda _{0}$ at 240 MHz) and 18 g (including the matching network), respectively. Each miniaturized antenna can only provide 2.4 MHz (~1%) bandwidth and $25~Omega $ input impedance. Having proper electromagnetic coupling between a cluster of three of such elements, measurement results demonstrate that the middle element can provide 18.4 MHz bandwidth (7.7%, 7.7 fold improvement) and $126~Omega $ input impedance when the other two elements are loaded with optimal reactive elements. Moreover, the proposed configuration shows 1 dB gain improvement at boresight direction. The effects of flight formation fluctuations on the coupled antennas are also investigated, and it is shown that as the separation distance between the coupled antennas increases, the input impedance and bandwidth enhancement decrease. To overcome this problem, a tunable matching circuit containing a single varactor is proposed. Measurement results are shown to be in good agreement with the simula-
ion predictions.]]>664172517352361<![CDATA[The Observable Field for Antennas in Reception]]>664173617461787<![CDATA[Linearly Polarized and Circularly Polarized Wideband Dipole Antennas With Reconfigurable Beam Direction]]>$0.51lambda _{0}times 0.51lambda _{0}times 0.39lambda _{0}$ for the LP antenna and $0.95lambda _{0}times 0.95lambda _{0}times 0.46lambda _{0}$ for the CP antenna, where $lambda _{0}$ represents the free-space wavelength at the center frequency of 2.2 GHz.]]>664174717553346<![CDATA[Improved Fruit-Fly Optimization Algorithm and Its Applications in Antenna Arrays Synthesis]]>664175617662619<![CDATA[Wideband Wide-Scanning Phased Array With Connected Backed Cavities and Parasitic Striplines]]>${le} 2$ . To compose the array, the open-backed cavities of adjacent antenna elements are connected rather than isolated to each other. The infinite array unit cell achieves about 43.4% bandwidth (7.4–11.5 GHz) with active VSWR ${le} 2$ when scanning up to ±60°, and nearly 44% bandwidth (7.45–11.65 GHz) with VSWR ${le} 2.5$ when scanning up to ±70° in all azimuth planes. For verification purpose, a $2times18$ array prototype is fabricated and it experimentally achieves nearly 51% bandwidth (7.2–12.1 GHz) with VSWR ${le} 2.3$ as scanning to 60 °, and more than 54% bandwidth (7.2–12.5 GHz) with VSWR ${le} 3.2$ up to 70° in the H-plane, showing close agreements with simulations.]]>664176717756642<![CDATA[Chip-to-Chip Switched Beam 60 GHz Circular Patch Planar Antenna Array and Pattern Considerations]]>$2{times} 2$ grid arrangement forming a planar array. The array is designed to fit on a typical multicore chip for reconfigurable interchip wireless communication. The array main beam is switched by changing the interelement phase shifts in the azimuth plane. The switching of the main beam is analyzed and verified through full-wave simulation. The design presented is an improvement over a previous design of a two-element antenna array. The Friis transmission equation with polarization components taken into account is used to model the interchip wireless link. To verify the model, a transmission coefficient measurement is made between a pair of the two-element arrays separated by a 10 mm distance. Both simulated and measured radiation patterns of the two-element array are presented for use in the Friis equation to calculate the transmission coefficients. Full-wave simulation of the array pair is also performed. The calculated results obtained from the Friis model agree well with both the measured and full-wave simulation results. The Friis model is used to calculate both signal and interference levels.]]>664177617874823<![CDATA[Compact Orientation Insensitive Dipolar Patch for Metal-Mountable UHF RFID Tag Design]]>$times30$ mm $times1.6$ mm ($0.0915lambda times 0.0915lambda times 0.0049lambda $ ), is proposed for designing a metal mountable tag. It consists of two pairs of orthogonal dipolar patches, which are placed in such a way that their inherent blind spots and null points are all removed. The tag antenna is readable in all directions in the boresight. Highly inductive thin stubs are inserted into the patches, and they are stub shorted to ground for increasing the tuning range of the operating frequency. A simple equivalent circuit model has been derived for visualizing the tag impedance characteristics. The proposed tag antenna can be read in the range of 2.9–3.5 m (at effective isotropic radiated power of 4 W) covering the entire azimuth plane when mounted on a metal plate with a dimension 20 cm $times20$ cm. It can also be read from at least 2 m omnidirectionally when used for dielectrics with $varepsilon _{r}$ in the range of 1–12.]]>664178817952417<![CDATA[Circular Polarization and Mode Reconfigurable Wideband Orbital Angular Momentum Patch Array Antenna]]>$2 times 2$ patch array. By changing the dc bias voltages of p-i-n didoes, the operating state of the antenna can be reconfigured between the left-hand CP with a mode $l = +1$ and right-hand CP (RHCP) with a mode $l = -1$ , over the wide impedance bandwidth of 21%. For the two states, the measured peak gains are larger than 5.3 and 5.2 dBi, and the axial ratios in half-power main beam directions are lower than 1.29 and 1.05 dB, respectively. Different from existing OAM antennas, the diversities of polarization, frequency, and mode, as well as the CP with inherently high cross-polarization discrimination are simultaneously integrated in the proposed antenna, which serves as a good basis for improving the capacity and spectral efficiency of radio systems. In addition, the antenna features single-external input port, a simple RFN occupying a small number of diodes, and a compact size of $1.28 times 1.28 times 0.07 lambda _{0}^{3}$ ($lambda _{0}$ is the free-space wavelength at center frequency), all of which ease implementation and reduce fabrication cost. Therefore, the proposed antenna is very promising for modern high-speed and large capacity wireless communication systems.]]>664179618042088<![CDATA[Fast and Accurate Simulation Technique for Large Irregular Arrays]]>664180518174479<![CDATA[A 1.2–12 GHz Sliced Notch Antenna Array]]>$19 times 19$ arrays—the proposed SNA and its Vivaldi counterpart—each operating over a 1.2–12 GHz (10:1) band. Both arrays are built using typical vertically integrated printed-circuit board cards, and are designed to exhibit VSWR < 2.5 within a 60° scan cone over most of the 10:1 band as infinite arrays. Measurement results compare very favorably with full-wave finite array simulations that include array truncation effects. The SNA array element demonstrates well-behaved polarization performance versus frequency, with more than 20 dB of D-plane $theta !=!45 {^{circ }}$ polarization purity improvement at the high frequency. Moreover, the SNA element also: 1) offers better suppression of classical Vivaldi E-plane scan blindnesses; 2) requires fewer plated through vias for stripline-based designs; and 3) allows relaxed adjacent element electrical contact requirements for dual-polarized arrangements.]]>664181818263007<![CDATA[Low Cross-Polarization Vivaldi Arrays]]>664182718374529<![CDATA[Backward Reflection in the Fresnel Problem]]>664183818451035<![CDATA[Extraordinary Transmission Through a Single Dielectric-Loaded Slot in a Thick Metallic Shield]]>664184618531264<![CDATA[An Improved Multifunctional Active Frequency Selective Surface]]>664185418624758<![CDATA[Computation of Tensor Green’s Functions in Uniaxial Planar-Stratified Media With a Rescaled Equivalent Boundary Approach]]>664186318732511<![CDATA[Broadband and Broad-Angle Multilayer Polarizer Based on Hybrid Optimization Algorithm for Low-Cost Ka-Band Applications]]>664187418812684<![CDATA[Simulation of High-Power Microwave Air Breakdown Modeled by a Coupled Maxwell–Euler System With a Non-Maxwellian EEDF]]>664188218936531<![CDATA[A Method of Suppressing Higher Order Modes for Improving Radiation Performance of Metasurface Multiport Antennas Using Characteristic Mode Analysis]]>664189419026168<![CDATA[Internally Combined Volume-Surface Integral Equation for EM Analysis of Inhomogeneous Negative Permittivity Plasma Scatterers]]>664190319132711<![CDATA[A Reverse Operation Self-Consistent Evaluation Approach for Singular Integrals in the SIE Analysis of PEC Targets]]>664191419242930<![CDATA[All-Analytical Evaluation of the Singular Integrals Involved in the Method of Moments]]>664192519361160<![CDATA[A Discontinuous Galerkin Surface Integral Equation Method for Scattering From Multiscale Homogeneous Objects]]>664193719463002<![CDATA[Solutions for General-Purpose Electromagnetic Problems Using the Random Auxiliary Sources Method]]>664194719564983<![CDATA[Formulation and Iso-Geometric Analysis of Scalar Integral Equations for Electromagnetic Scattering]]>664195719661837<![CDATA[Self-Interference Channel Characterization for Wideband $2times 2$ MIMO Full-Duplex Transceivers Using Dual-Polarized Antennas]]>$2times 2$ multi-input multi-output full-duplex transceiver using dual-polarized antennas. The measured SI channel power delay profiles at 2.45 GHz with 500 MHz span in various environments, anechoic chamber, laboratory room, and corridor, reveal that the SI channel can be represented by a multipath model consisting of two components: a quasi-static internal SI subchannel due to the specific Tx/Rx antenna structure and a time-varying external SI subchannel due to possible reflections from the surrounding environment. The quasi-static internal SI subchannel parameters can be derived from the Tx/Rx antenna structure specifications. The time-varying external SI channel exhibits cluster arrival features and can be represented by a modified Saleh–Valenzuela model with lognormal-distributed taps, and the cluster power exponentially decays with cluster arrival delay. However, the path power-versus-arrival-delay decay is exponential in a laboratory-room environment while it follows the power law in a corridor environment. The SI channel coherence bandwidth and delay spread in all three measurement environments follow normal and lognormal distributions, respectively.]]>664196719763762<![CDATA[Magnitude and Spectrum of Electromagnetic Wave Scattered by Small Quadcopter in $X$ -Band]]>2 obtained by numerical simulation and direct measurements in an anechoic chamber at a frequency of 9 GHz and their comparison are discussed. Effectiveness of the parasitic leakage suppression technique at small RCS measurements has been shown.]]>664197719843859<![CDATA[A Discrete RET Model for Millimeter-Wave Propagation Through Vegetation]]>664198519983458<![CDATA[Level Crossing Rate and Average Fade Distance of Signal Scattering From Rough Layered Interfaces]]>664199920071469<![CDATA[1.5-D Sparse Array for Millimeter-Wave Imaging Based on Compressive Sensing Techniques]]>664200820152954<![CDATA[A Linear Method for Shape Reconstruction Based on the Generalized Multiple Measurement Vectors Model]]>664201620252427<![CDATA[A Simple Approach for Estimating the Effective Electric Parameters of 2-D Targets]]>virtual experiments concept, as it exploits a combination of the incident fields to be able to focus the exciting field on a certain region. Then, it associates the effective parameters to the permittivity and conductivity values that minimize the mean square error between the virtual scattered field (obtained by recombining the measured data) and the field scattered by a homogeneous dielectric cylinder centered on the target, having approximately the same area. The center of this effective scatterer and its radius can be estimated with a qualitative imaging algorithm or can be given as a priori information. Notably, by properly designing the virtual incident fields, the proposed method can easily handle the case of multiple targets. The effectiveness of the method is assessed by both numerical and experimental examples.]]>664202620342006<![CDATA[High-Order Sparse Shape Imaging of PEC and Dielectric Targets Using TE Polarized Fields]]>664203520432733<![CDATA[Correcting Gravitational Deformation at the Tianma Radio Telescope]]>$mu text{m}$ , and the repeatability is sufficient. We have acquired a model for the gravitational deformation of the TMRT. After applying the model, there is a 150%–400% improvement in the aperture efficiency at low and high elevations. The model flattens the gain curve between 15°–80° elevations with an aperture efficiency of approximately 52%. The final weighted root-mean-square error is approximately 270 $mu text{m}$ . The e-OOF technique reduces the constraints on the telescopes.]]>664204420481070<![CDATA[Design and Optimization of Multiport Pixel Antennas]]>664204920541425<![CDATA[A Wideband Differentially Fed Dual-Polarized Stacked Patch Antenna With Tuned Slot Excitations]]>$Gamma $ -shaped feeding line. To connect to the differential circuit easily, a rotated feeding mechanism is adopted. A prototype of the proposed antenna is fabricated and tested. Measured results show that the antenna has a wide differential impedance bandwidth of 49.4% (1.66 to 2.75 GHz) for $vert text{S}_{mathrm{ dd11}}vert ~f-15 $ dB. In addition, a high differential port isolation of more than 37 dB, a stable radiation pattern with a 3 dB beamwidth of 61°±5°, and a stable gain around 8.7 dBi are obtained within the operational band. The proposed antenna features wideband, high port isolation, and stable gains, and it is a promising candidate for base station systems using differential signals.]]>664205520602508<![CDATA[Fabry–Pérot Resonator Antenna With High Aperture Efficiency Using a Double-Layer Nonuniform Superstrate]]>$3.5~lambda $ (where $lambda $ is the wavelength in free space) is presented. In comparison with a uniform superstrate, the proposed DNS enhances the FPRA’s directivity from 19.6 to 20.4 dBi and correspondingly improves the aperture efficiency from 76.3% to 91.7%.]]>664206120661954<![CDATA[Inverted-Sigmoid Shaped Multiband Dielectric Resonator Antenna With Dual-Band Circular Polarization]]>664206720721836<![CDATA[A Circularly Polarized Octagon-Star-Shaped Microstrip Patch Antenna With Conical Radiation Pattern]]>11 modes from the two superimposed square patches, omnidirectional CP radiation is achieved for the proposed antenna. The operation mechanism of the antenna is theoretically explained. A prototype of the proposed antenna is fabricated and measured. Measured results are consistent with the simulated and theoretical results. The antenna can produce a conical CP radiation pattern with a measured average axial ratio (AR) of 1.80 dB in the azimuth plane ($theta = 45$ °) and a measured 3-dB AR bandwidth covering the global positioning system L1 band.]]>664207320783071<![CDATA[Doubly Curved Reflector Design for Hybrid Array Fed Reflector Antennas]]>664207920831106<![CDATA[In-Band Full-Duplex Multimode Lens-Loaded Eight-Arm Spiral Antenna]]>$4 times 4$ Butler matrix beamforming networks (BFNs) to differentiate between the TX and RX arms/ports and distinguish their functionalities. In the absence of BFN imbalances and antenna geometry asymmetries, the isolation is theoretically infinite due to full cancellation at the RX antenna ports and RX beamformer ports. This approach also enables diverse co-polarized TX and RX radiation modes while maintaining theoretically infinite isolation. The considered spiral aperture modes are mode 1, with a broadside pattern, and modes 2 and 3, with conical patterns. The aperture is loaded with a hyperhemispherical dielectric lens to improve the far-field performance. The operational principles are discussed first under ideal conditions, followed by the computational and experimental results. Average measured isolation >38 dB is achieved over 3:1, 1.5:1, and 1.3:1 bandwidths for the radiating TX and RX modes 1–3, respectively. Once the TX and RX are driven with identical modes, similar radiation patterns with high envelope correlation coefficient are obtained.]]>664208420897009<![CDATA[Wideband Reconfigurable Crossed-Dipole Antenna With Quad-Polarization Diversity]]>664209020941670<![CDATA[Transparent Patch Antenna Using Metal Mesh]]>$20~mu text{m}$ ) was used to fabricate transparent patch antennas. The realized gain, efficiency, front-to-back ratio, and radiation patterns of the transparent MM patch antennas are discussed for various combinations of radiator and ground planes. Overall, the transparent MM patch antennas offer optical transparency, maintain sufficient performance, and could possibly be used in wireless transparent applications.]]>664209521002319<![CDATA[Upper Bounds on the Bandwidth of Electrically Small Single-Resonant UHF-RFID Tags]]>$surd 2$ times (approximately 41%) higher, requiring a proper relaxation of the matching level at resonance. It is also shown that bandwidth of small real tags with perfect matching, which is typically far from its upper bound, can also be enhanced approximately the same factor at the expense of a small reduction (13.4%) in the peak read range. A practical example is provided, where two small split-ring resonator-based tags of the same size ($k_{0}a=0.31$ ) are designed. It demonstrates that such improvement on the tag bandwidth can be approximately obtained by simply changing the chip position, without the need of an external matching network. The improved tag was fabricated and measured, as a proof of concept. The results obtained from the proposed analysis allow RFID designers to determine how well a tag performs, compared to theoretical bandwidth limits.]]>664210121061558<![CDATA[A Novel Feeding Network for Quadri-Polarization Antennas With Simultaneously Wide Impedance and Polarization Bandwidth]]>664210721122709<![CDATA[Dual-Band Fresnel Zone Plate Antenna With Independently Steerable Beams]]>664211321182812<![CDATA[A Conformal Circularly Polarized Antenna for Wireless Capsule Endoscope Systems]]>664211921241983<![CDATA[Design of Filtering-Radiating Patch Antennas With Tunable Radiation Nulls for High Selectivity]]>664212521301721<![CDATA[Stochastic Polynomial Chaos Expansion Analysis of a Split-Ring Resonator at Terahertz Frequencies]]>66421312134694<![CDATA[Microstrip Colinear Antenna Array for a Small Unmanned Aerial System Lobe Differencing Correlation Radiometer]]>$2times 2$ rectangular L-band microstrip array used for the CU-BlackSwift Technologys (BST) lobe-differencing correlation radiometer is presented. The antenna array was integrated into the fuselage of the BST tempest small unmanned aerial system (sUAS). A prototype $2times 1$ microstrip colinear (MiCo) antenna array operating at center frequency 1.4135 GHz was characterized to validate the accuracy of HFSS simulation results of input impedance. Furthermore, a $2times 2$ MiCo antenna array with effective use of styrofoam blocks for frequency tuning and providing mechanical stability was built. The effect of separation of the upper and lower array pairs on mutual coupling and radiometer sensitivity and the tuning effect of the styrofoam blocks on resonant frequency were studied. An optimal vertical separation distance was identified that minimizes mutual coupling and provides maximum main-to-back lobe ratio. A method of antenna frequency tuning using dielectric overlay material is presented to correct the dielectric impact of the sUAS fuselage.]]>664213521401674<![CDATA[A Direct Proof of SFIE]]>66421412142223<![CDATA[Controlling Leaky Waves With 1-D Cascaded Metasurfaces]]>66421432146507<![CDATA[An Operator Absorbing Boundary Condition for the Absorption of Electromagnetic Waves in Dispersive Media]]>66421472150798<![CDATA[A Combined Source Integral Equation With Weak Form Combined Source Condition]]>664215121551426<![CDATA[Two-Way Propagation Modeling of Expressway With Vehicles by Using the 3-D ADI-PE Method]]>664215621601203<![CDATA[Statistical Distribution of the Enhanced Backscatter Coefficient in Reverberation Chamber]]>RC, the analytical expressions for the statistical distributions of the enhanced backscatter coefficient (${e}_{b}$ ) are derived. It has been found that the superposition of random variables with exponential distributions has a probability density function of Erlang distribution, and the distribution of the ratio of two random variables with Erlang distribution can be derived analytically, which is the distribution of the enhanced backscatter coefficient. The unbiased estimator of ${e}_{b}$ is also given with and without approximations. Measurements have been performed to verify the results.]]>664216121641204<![CDATA[Statistical Evaluation of the Azimuth and Elevation Angles Seen at the Output of the Receiving Antenna]]>66421652169924<![CDATA[Scalp-Implantable Antenna Systems for Intracranial Pressure Monitoring]]>3 (System A) and 406 mm^{3} (System B). Each system is integrated with the microelectronic components and a battery. The key feature of the proposed implantable antenna is its small volume ($8 times 6 times 0.5$ = 24 mm^{3}) with a slotless and a vialess ground plane, thereby reducing design complexity. Moreover, the structure exhibits satisfactory peak gain values of −28.5 and −22.8 dBi at a lower and higher resonant band, respectively. Simulations using the finite-element method and finite-difference time domain are performed to evaluate the implantable systems. For validation, measurements are carried out in a saline solution. The antenna offers a good agreement between the measured and simulation results. In addition, a link budget is calculated to analyze the data transmission range.]]>664217021731656<![CDATA[Member Get-A-Member (MGM) Program]]>664217421743456<![CDATA[Fuel Your Imagination]]>664217521751718<![CDATA[Introducing IEEE Collabratec]]>664217621762193<![CDATA[IEEE Transactions on Antennas and Propagation]]>664C3C3112<![CDATA[Institutional Listings]]>664C4C4389