A WiFi and Bluetooth Low-Energy Backscatter Combo Chip With Beam Steering Capabilities

This article introduces a dual-mode backscatter integrated circuit that supports both WiFi and Bluetooth low-energy (BLE) transmissions. It enables a multiantenna WiFi mode with reconfigurable beam steering of single-sideband (SSB) quadrature phase shift-keying (QPSK) signals, while also facilitating omnidirectional SSB BLE-to-BLE backscatter communication. To achieve beam steering, two techniques are proposed: 1) a transmission-line-less fully reflective SP4T backscatter switch is employed to minimize power loss and maximize the communication range and 2) a multiantenna array is constructed using the aforementioned SP4T switches together with a baseband phase-shifting technique to reradiate the incident WiFi signal with a controllable angle of direction. The chip implementation is based on a 65-nm CMOS process and operates at a power consumption of <inline-formula> <tex-math notation="LaTeX">$5.5 \mu \text{W}$ </tex-math></inline-formula> in standby mode. In backscattering mode, it consumes <inline-formula> <tex-math notation="LaTeX">$39 \mu \text{W}$ </tex-math></inline-formula> for the single-antenna approach and <inline-formula> <tex-math notation="LaTeX">$88 \mu \text{W}$ </tex-math></inline-formula> for the multiantenna approach. The proposed design achieves a worst-case access point (AP)-to-AP range of 35 and 56 m for the single-antenna and multiantenna approaches, respectively.


I. INTRODUCTION
T HE RAPID growth of the Internet of Things (IoT) has led to an increasing demand for low-cost and low-power connectivity solutions.To keep costs low, it makes sense to leverage already extensively deployed infrastructure: in the case of IoT devices, many already use WiFi and/or Bluetooth low energy (BLE).WiFi is a well-established wireless networking technology that provides high-speed data transfer and relatively widespread local coverage, while BLE, on the other hand, is specifically designed for energyefficient communication over short distances.However, the inclusion of WiFi/BLE transceivers in IoT devices inevitably leads to high power consumption because of the operation of active RF circuit blocks: most WiFi transceivers require 10-100 mW of power, while most BLE transceivers require 1-10 mW.These levels of power consumption can constrain the size and lifetime of IoT devices, as they require either a continuous connection to a power outlet or large batteries.This may not be practical or convenient, especially for devices deployed in remote or inaccessible locations.
To address this challenge, backscatter techniques, drawing inspiration from RFID technology, have been developed as an ultralow-power solution for generating WiFi/BLE signals without requiring active RF circuit blocks on the IoT devices.For example, several studies have explored the generation of WiFi 802.11b signals from an incident tone [1], [2], [3].However, these approaches require additional tone generators that are not readily available in commodity hardware, thereby increasing the total cost of system deployment.Alternatively, recent research has introduced ambient backscatter, which capitalizes on existing WiFi/BLE signals [4], [5], [6], [7], [8].By modulating the phase of the incident transmitted signal, such approaches eliminate the need for dedicated tone generators and optimize the utilization of existing infrastructure.
Nonetheless, the passive nature of ambient backscatter imposes limitations on communication range, resulting in increased deployment costs as more access points (APs) may need to be added.To understand this limitation, it is important to consider the link budget, which accounts for path loss in the air and insertion loss on the tag, as the signal travels from the transmitter to the tag and back to the receiver, as shown in Fig. 1(a).Transmitted power from the transmitter is constrained by FCC regulations-which are fixed and cannot be changed-and receiver sensitivity is fundamentally dictated by noise and bandwidth requirements-and commercial receivers have optimized these already and further improvements are not practically possible.Consequently, the range can only be improved by either reducing insertion loss on the tag or employing multiantenna gain (MIMO gain).
Previous work [7] [Fig.1(b)] proposes a fully reflective backscatter reflector to minimize insertion loss and a Van-Atta-array-based retro-reflective backscatter array to enhance the range.However, bulky transmission lines used to create the quadrature phase difference between antennas occupy large PCB space and lack tunability.Additionally, the retro-reflective approach necessitates co-located transmitter and receiver units that must be full-duplex, or else self-interference may degrade the receiver's functionality.Regrettably, there are presently no commercial WiFi/BLE full-duplex transceivers available, even when considering different receiving and transmitting frequencies.
To overcome the limitations imposed by the requirement for co-located transmitter and receiver, and to optimize range while leveraging MIMO gain, this article introduces a beamsteering approach that utilizes beam-forming techniques without the use of power-intensive [9] RF phase shifters.Instead, phase shifts are absorbed into the intermediate frequency (IF) domain on the tag, resulting in a total power consumption of only 88 μW.Simultaneously, the proposed SP4T reflector ensures that the beam-steering backscatter system achieves MIMO gain while minimizing power loss during signal reflection.
As illustrated in Fig. 1(c), the proposed beam-steering system offers a compact and flexible steerable beam solution that eliminates the need for bulky transmission lines, making it suitable for integration into mesh networks.While MIMO has been explored in various wireless systems, the range improvement achievable through beam-forming from a single tag has received limited attention.For instance, previous works, such as VMScatter [10], have focused on increasing throughput and reducing bit error rate (BER) through spacetime coding mechanisms.MOXcatter [11] implements an antenna array on the tag but cannot fully exploit the benefits of a MIMO setup due to clocks with identical phases driving the backscatter switches.On the other hand, PushID [12] utilizes distributed MIMO readers to create energy peaks in space.This approach eliminates the need to fine-tune optimal beam-forming parameters at RFID tags prior to communication.However, implementing such a distributed MIMO reader system requires a higher number of readers, leading to increased deployment costs.
In addition to addressing the range limitation, this work also fills a gap in prior research by presenting a BLE backscatter solution specifically designed for integrated circuits, which is crucial considering that WiFi may not always be available in low-power applications.
The initial presentation of this combo-chip solution was given in [13], whereas this article provides a more comprehensive analysis, system design considerations, and implementation details.Section II introduces the fully reflective transmission-line-less backscatter and the beam-steering technique.Section III explores the concept of BLE backscatter, followed by a detailed discussion of the combo-tag implementation in Section IV.Measurement results are presented in Section V, and finally, Section VI concludes this article.

II. FULLY REFLECTIVE BEAM-STEERING TECHNIQUES A. PROPOSED TECHNIQUE 1: FULLY REFLECTIVE TRANSMISSION-LINE-LESS BACKSCATTER APPROACH
Fig. 2(a) showcases the single-sideband (SSB) architecture proposed in [6].In this architecture, the incident RF signal traverses a Wilkinson power splitter, splitting the power equally between two branches.To achieve SSB backscatter, the two branches are controlled by IF clocks (IF I and IF Q ) with a 90-degree phase offset.When IF I is low, the signal in the upper branch is absorbed by the 50-load, while it is fully reflected by the open load when IF I is high.Similarly, when IF Q is low, the signal in the lower branch is dissipated by the 50-load, and it is reflected with a 90-degree phase shift by the 1.2-pF capacitor load when IF Q is high.In order to compute the insertion loss of this approach, we observe that within one IF clock duration, T 0 , four different load combinations are cycled through.Taking advantage of this periodicity, we can analyze the Fourier series within a single IF clock period.However, to proceed, we first need to determine the equivalent load for each state.Fig. 2(a) illustrates that when the open load and 50-load are connected, the equivalent reflection coefficient seen from the antenna is 1 = −0.5.The other cases are also listed in the figure.When a unit magnitude incident wave of carrier frequency f c enters the antenna, the reflected signal can be represented as sin(2π f c t) × S R , where S R is a periodic function of the IF frequency.The amplitude and phases of S R are determined by the load condition and can be expressed as S R = ∞ n=−∞ c n e j2π nf 0 t , where f 0 = 1/T 0 .Since the backscatter signal is dominated by the first harmonic, the coefficient for n = 1 matters, and can be calculated by The calculated insertion loss, based on the magnitude of the first harmonic coefficient c 1 , is determined to be 9.94 dB, considering the given reflection coefficients 1 -4 .To better understand the factors contributing to this loss, three main elements were identified: 1) the 50-load absorbing half of the incident power during the IF period; 2) power loss due to the splitting and combining actions of the Wilkinson power splitter; and 3) the presence of odd harmonic backscatter images apart from the first harmonic component.
In an effort to mitigate the loss caused by the 50load, Meng et al. [7] replaced it with a short and inductor configuration, as depicted in Fig. 2(b).The equivalent reflection coefficients from the antenna were determined to be 0.707∠ − 135 • , 0.707∠135 • , 0.707∠45 • , and 0.707∠ − 45 • .By applying the same method to calculate the first harmonic coefficient, the insertion loss was reduced to 3.94 dB, primarily attributed to the Wilkinson power splitter and other irrelevant harmonic components.
Previous attempts to improve the insertion loss involved selecting different loads; however, power splitting and combining inherently result in power loss.An alternative approach, demonstrated in Fig. 2(c), is to combine the I/Q path at the IF stage instead of using the Wilkinson splitter for I/Q combination at RF.By employing an SP4T switch with appropriate termination at each branch controlled by a 25% duty cycle IF clock, the same SSB backscatter operation can be achieved, as the four states still cycle through within one IF clock period.The terminations at each branch are specifically chosen as follows: short ( = e j180 • ), open ( = e j0 • ), 1.2-pF capacitor ( = e −j90 • ), and 3.3-nH offchip inductor ( = e j90 • ), in order to maximize the reflected power.An off-chip inductor is chosen to enable post-silicon trimming and to achieve a higher Q.By employing (1) to calculate the first harmonic coefficient, the insertion loss is further reduced to 0.94 dB, representing the theoretical minimum loss achievable through the use of hard-driving switches.This is the approach used throughout this article.

B. PROPOSED TECHNIQUE 2: PHASED-CONTROLLED BEAMSCATTER APPROACH
Analog beamforming plays an increasingly crucial role in multiple-input-multiple-output communication systems.The traditional approach, illustrated in Fig. 3(a), involves utilizing digital controlled phase shifters to manipulate the relative phase between elements in the antenna array.This manipulation allows for the steering of the beam toward a specific direction, resulting in a longer and narrower main lobe radiation pattern.In order to optimize performance, such as maximizing received power, the digital signal processing block determines the optimal phase shifter weights to control.However, conventional active phase shifters encounter challenges, such as loss, nonlinearity, and noise, which can limit their effectiveness.Additionally, passive phase shifters, although an alternative, often require significant space and can contribute to larger form factors [9].
When the antennas are separated by distance L, by controlling the relative phase , the following formula can be used to estimate the direction of the beam: where θ is the steering angle and λ is the wavelength.For example, when = 30 • , the beam can be steered by 10 • when the antenna separation is set as half wavelength.A separation distance of 6.25 cm is used in this article for the 2.4-GHz ISM band unless otherwise specified.
In order to capitalize on the advantages of MIMO gain without the use of lossy or power-consuming phase shifters, [7] introduces the implementation of the Van Atta array [14] to retro-reflect the backscattered beam back to the transmitter.This is accomplished by employing two pairs of antennas interconnected with a quadrature-phase shifted transmission line, as depicted in Fig. 3(b).This configuration enables quadrature phase shift-keying (QPSK) SSB backscatter without the need for power-absorbing components like Wilkinson power splitters and 50-resistors, resulting in a noteworthy 12-dB gain improvement compared to the singleantenna approach shown in Fig. 2(b).Nevertheless, it is important to note that transmission lines occupy considerable PCB area, operate at specific frequencies, and possess limited phase tunability.Furthermore, since the beam is consistently steered back to the transmitter, co-located APs become a requirement, which can introduce self-interference issues.Alternatively, to achieve full control over the beam direction, the backscatter switch array can be connected to RF phase shifters, allowing the reflected signal to be combined in a manner similar to traditional beamforming systems, as depicted in Fig. 3(c).The backscatter IF clock generator serves the IF clocks in the same manner as the single-antenna approach, with the clocks being directed to the duplicated reflectors.Phase shifters are inserted after each reflector to enable beamforming.However, this approach shares the downside of conventional beamforming, as it requires highpower active RF phase shifters or lossy passive RF phase shifters.
To mitigate the need for RF phase shifters, the proposed beam-steering backscatter configuration in Fig. 3(d) moves the phase shift to the IF domain.In this setup, the reflector unit in the array is driven by IF clocks with a delay compared to the neighboring unit, effectively absorbing the phase shifts into the IF domain.This design achieves lowpower operation as the complexity resides in the IF domain, which operates at tens of MHz instead of at GHz.

III. BLE BACKSCATTER
In scenarios where WiFi is unavailable, such as with some wearable devices, BLE emerges as an ideal option for backscattering, particularly for transmitting small amounts of data.As illustrated in Fig. 4(a), BLE operates in the 2.4-GHz ISM band and encompasses 40 channels spaced by 2 MHz, with a center frequencies ranging from 2402 to 2480 MHz.
Unlike WiFi, which employs phase modulation, BLE utilizes GFSK modulation.The 1-Mb/s data stream is represented by one of two modulation frequencies: symbol 1 is represented by a positive frequency deviation of +250 kHz, while symbol 0 is represented by a negative frequency deviation of -250 kHz from the channel's center frequency.A Gaussian filter is employed to shape the deviation before transmission.For establishing connections, BLE devices transmit advertising packets across three advertisement channels: channel 37 (2402 MHz), channel 38 (2426 MHz), and channel 39 (2480 MHz).
To enable backscattering, RBLE [15] leverages the concept from [5] by transforming the incident signal into a single tone through reverse-whitening, where the data is set to all 0s or 1s.A direct frequency shift scheme based on binary frequency shift keying (BFSK) is utilized.For instance, in Fig. 4(b), if the input signal is set to symbol 0 at channel 37, and the target backscatter channel is channel 3, when the tag data is 1, the backscatter frequency will be on the upper side of channel 3, resulting in an IF of 8.5 MHz.Conversely, when the tag data is 0, the backscatter frequency will be on the lower side of channel 3, with an IF equal to 8 MHz.
Another backscattering method, Freerider [16], is shown in Fig. 4(c), which modulates tag data 1 by changing the BLE symbol 0 to 1 or 1 to 0, while tag data 0 keeps the BLE symbol unchanged.This modulation takes place in the same channel as the incident signal, inevitably producing an image copy within the channel.Unlike WiFi, where single-sideband cancellation can be employed, BLE tags are unaware of the incident symbols and are thus unable to determine which sideband should be canceled.
Fig. 4(d) presents the proposed approach to address the issue of the backscattered signal landing in the incident channel, as observed in [16].This is achieved through IF frequency translation during backscatter.Taking the translation from Channel 38 to Channel 37 as an example, when the tag data is 0, the backscattered signal remains unchanged, resulting in an IF of 24 MHz.On the other hand, when the tag data is 1, to move symbol 0 in Channel 38 to symbol 1 in Channel 37, an IF of 23.5 MHz is required, while moving symbol 1 in Channel 38 to symbol 0 in Channel 37 necessitates a frequency of 24.5 MHz.However, since the incident signal cannot be controlled without knowledge of its content, it is necessary to generate these two possibilities simultaneously.This can be achieved by XOR-ing a 0.5-MHz clock with a 24-MHz clock.The resulting 1-MHz away image remains the same as in [16], but it can be easily eliminated by employing a digital filter at the receiver.This is the approach taken in this work.
It is worth mentioning that PBLE [17] provides an alternative method of converting the tone to a BLE signal by modulating the phase of the incident signal instead of directly shifting its frequency, assuming BFSK.This approach relies on the fact that instantaneous frequency change can be equivalently converted to instantaneous phase change.Given the BLE GFSK modulation index of 0.5, the phase shift between each symbol is always either plus or minus 90 • , enabling the generation of IF clocks using a quadrature clock with a state machine.This alternative approach yields an improved packet error rate (PER) compared to the BFSK approach, though is not implemented in this work.

IV. TAG IMPLEMENTATION A. SYSTEM OVERVIEW
The block diagram of the designed backscatter tag is depicted in Fig. 5.It incorporates an envelope-detector-based wake-up receiver, designed to detect a specific WiFi wake-up pattern with programmed packet lengths and gaps [6], [7].The hierarchical wake-up scheme, extensively described in [18], provides further insights into this process.The wake-up receiver's baseband clock, denoted as CLK Program , is generated by dividing an off-chip 16-MHz crystal.Upon receiving the wake-up signal, a phase-locked loop (PLL) is activated and generates multiple-phase IF clocks utilizing a 2-MHz reference clock, which is also obtained by dividing the 16-MHz crystal.
The SP4T reflector control signals, namely, IF OUT,IQ , are generated by a single-antenna WiFi/BLE modulator that combines the quadrature IF clocks with 2-bit tag data.A beam-steering modulator digitally phase shifts the IF control signals, producing IF OUT,IQ , IF OUT,IQ + , IF OUT,IQ +2 , and IF OUT,IQ + 3 .These signals are then directed to the SP4T reflector arrays to complete the beam-forming process.In the subsequent sections, we will delve into a more detailed description of each of the circuit blocks involved in the system.

B. PLL AND MULTIPHASE GENERATOR
To enable backscattering of WiFi signals from Channel 6 to either Channel 1 or Channel 11, a 25-MHz IF is required.Similarly, for backscattering BLE signals between Channel 37 and Channel 38, a 24-MHz IF is needed.To generate these frequencies, a standard type-II integer PLL is employed, as illustrated in Fig. 6(a), leveraging a flexible divider ratio [19].
The 2-MHz reference clock is obtained by dividing the 16-MHz crystal oscillator.A 6-stage differential ring voltagecontrolled oscillator (VCO) is utilized to generate 12-phase clocks operating at either 48 MHz or 50 MHz.These clocks are subsequently fed through six divider-by-2 blocks, resulting in clocks with a 15-degree resolution at 24 MHz or 25 MHz.This configuration allows for the generation of the required IF frequencies for WiFi and BLE backscattering operations.

C. SINGLE-ANTENNA AND BEAM-STEERING MODULATOR
In the BLE-only implementation, a one-bit multiplexer (MUX) controlled by the one-bit tag data is utilized, as depicted in Fig. 6(b, top).When the tag data is 0, the IF clocks are directly passed through, resulting in a backscattered signal that is a frequency-translated version of the incoming BLE signal.Conversely, when the tag data is 1, the MUX combines the IF clocks with a 0.5-MHz clock, resulting in a backscattered signal that is translated to the opposite FSK tone along with the image tone.
On the other hand, Fig. 6(b, bottom) illustrates the implementation for a WiFi/BLE combo-chip.In BLE mode, one control signal for the 2-bit MUX is disabled (grounded), and another control signal is connected to a one-bit MUX.The output of this MUX is the one-bit tag data when the tag data is 0, and the XOR of the tag data with 0.5 MHz when the tag data is 1.In WiFi mode, the two-bit MUX is controlled by the two-bit tag data to generate a QPSK SSB signal.
Fig. 6(c) illustrates the beam-steering control logic, which consists of three layers of clock selection trees.The first layer is the angle selection logic, responsible for determining the phase difference between each reflector.The second layer is the tag data selection logic, controlled by the 2-bit data, serving the same purpose as the single-antenna modulator shown in Fig. 6(b).Finally, the third layer is the plus-minus control logic, which determines whether the clock phases applied to each reflector are delayed or lead compared to the neighboring one.An identical circuit is implemented for the quadrature portion, but with all signals shifted by 90 • to generate the SSB signal.

V. MEASUREMENT RESULTS
The chip was manufactured in a 65-nm CMOS process, occupying a core area of 0.42 mm 2 as depicted in Fig. 7.The power breakdown is illustrated in Fig. 8.The wakeup receiver consumes 5.5 μW of power.In single-antenna WiFi/BLE mode, the total power consumption amounts to 39 μW, with the PLL module accounting for 30 μW.In beam-steering mode, the power usage of the modulator increases to 46 μW due to the complexity of the phaseshifting clock selection tree, resulting in a total power consumption of 88 μW.Fig. 9 displays the implemented PCB with the chip mounted on it.Four off-chip inductors are mounted on the back side of the board to avoid congestion with other signal traces on the top layer.Indoor wireless WiFi measurements were conducted using the setup depicted in Fig. 10(d) to capture the singleside-band backscatter spectrum.The spectrum analyzer was positioned in the direction where the main reflected beam was present to capture the maximum backscatter power.Simultaneously, the spectrum for the single-antenna case was also recorded for comparison.As shown in Fig. 10(a) and (b), the beam-steering approach achieved a  10-11 dB increase in backscatter power compared to the single-antenna approach, thanks to the utilization of MIMO gain.Additionally, both the lower-side-band and upper-sideband exhibited a strong 14-15 dB rejection of image tones.It is important to note that due to the close proximity between the transmitter (TX) and receiver (RX), the high-power input WiFi signal in channel 6 was consistently captured by the RX, contributing to the observed spectrum.
For wired BLE measurements, shown in Fig. 10(e), the single-side-band backscatter spectrum was obtained as depicted in Fig. 10(c).Due to limitations in circulator isolation and impedance mismatches, the input signal at channel 38 appeared in the received spectrum.Fig. 11(a) shows the setup to evaluate the beam-steering pattern of the backscattered signal.Here, a signal is transmitted normally to the tag from a transmitter, with the same phase at the input port of the antenna array by using a wired connection and a power splitter.A receiver, in this case, a spectrum analyzer, is positioned at a distance of 1 m from the tag.The reflected signal undergoes a frequency shift of 25 MHz and its phase is modified at each port of the array due to the distinct phases of the IF clocks controlling the switches.The reflected signals emerge from the antenna and combine to form a signal that represents the magnitude of the beam pattern, which is measured by the receiver.
To obtain different beam patterns, the backscatter tag is configured with specific -phase shifts between consecutive antennas and is rotated around its axis toward the receiver using a motor with fine angle control.The received signal power is then plotted as a function of the tag rotation angle.This process is repeated for various tag configurations, and the obtained results are compared with the theoretical beam pattern, as depicted in Fig. 11(b).A maximum gain of 11 dB is observed, which is close to the theoretical gain of 12 dB [calculated as 10× log( 16)] for a four-antenna array operating in backscattering mode.The slight deviation in the clock phases causes minor increases in side lobes within the measured beam patterns.With λ/2 antenna separation and 15 • of IF phase resolution, supported beam resolution is about 5 • , which is enough for demonstration purposes.However, the resolution requirement still depends on the range that certain applications need.tag, and both positions remain fixed throughout the entire measurement.The average transmitted power is 24 dBm with a 6-dB peak-to-average power ratio (PAPR) from the modulated signal.The tag is programmed with a specific beam-steering angle setting, and another COTS WiFi receiver is rotated around the tag to determine the maximum distance at which the receiver can decode the backscattered WiFi packets.Fig. 12(b) presents the results of the distance measurements for different beam-steering angles, specifically θ = −5 • , +5 • , −25 • , and +25 • .These results indicate that a maximum distance of 35 m can be achieved between the tag and the receiver when the receiver is positioned in the main beam direction, for a total distance between the COTS TX and RX is 58 m in this measurement.In the case of a single-antenna configuration, the tag-to-receiver distance remains constant at 13 m for all directions, given a tag-to-transmitter distance of 23 m and a COTS TX-to-RX distance of 36 m.Note that the total path loss is determined by the product of the TX-to-tag distance (D 1 ) and the tag-to-RX distance (D 2 ).In the  Fig. 12(c) compares the range achieved in this work with prior art approaches.In the single-antenna case, the worst-case range from the AP to the tag to the receiver is 35 m, which represents a 34% improvement compared to [7].In contrast, the MIMO case provides a worst-case range of 56 m from the AP to the tag to the receiver, exhibiting a 143% improvement over [7].The proposed beam-steering approach offers the advantage of maintaining communication in various angles, while [7] only supports co-located APs, limiting the maximum range to a single direction.
Fig. 13 shows the wireless BLE decoded results at a BER of 2 × 10 −4 .Bit repetition is used and the supported data date is 500 kb/s.Table 1 compares the proposed work with prior arts.This work shows the longest achievable range among state-of-theart backscatter systems and the first BLE backscatter system implemented in IC.

CONCLUSION
In this article, a WiFi/BLE combo backscatter chip is presented.Theoretical calculations for insertion loss are presented, demonstrating that the proposed power-splitterfree fully reflective modulator achieves the lowest theoretical insertion loss among different types of backscatter reflectors.An IF-based beam-steering technique is presented to address the self-interference issue coming from co-located APs, further increasing the range to a maximum of 56-m worst-case AP-to-AP distance.The IC consumes 39 μW and 88 μW for single-antenna and beam-steering approaches, respectively.The extended backscatter range benefits outdoor applications like smart doorbell security cameras or pet trackers, along with giving extra link margin to indoor IoT applications for either longer range or higher data rates.

FIGURE 4 .
FIGURE 4. Proposed BLE backscatter scheme.(a) BLE channels in 2.4GHz ISM band, (b) direct frequency shift backscattering scheme between two channels, (c) Freerider scheme which interchanges symbol while tag data is 1, and (d) proposed backscattering approach.

FIGURE 10 .
FIGURE 10.Measured and compared WiFi spectrum showing SSB backscatter-based frequency translation to (a) lower sideband and (b) upper sideband, and (c) BLE spectrum showing frequency translation from Ch.38 to Ch.37.(d) WiFi spectrum measurement setup, and (e) BLE spectrum measurement setup.

Fig. 12 (
Fig.11(a) shows the setup to evaluate the beam-steering pattern of the backscattered signal.Here, a signal is transmitted normally to the tag from a transmitter, with the same phase at the input port of the antenna array by using a wired connection and a power splitter.A receiver, in this case, a spectrum analyzer, is positioned at a distance of 1 m from the tag.The reflected signal undergoes a frequency shift of 25 MHz and its phase is modified at each port of the array due to the distinct phases of the IF clocks controlling the switches.The reflected signals emerge from the antenna and combine to form a signal that represents the magnitude of the beam pattern, which is measured by the receiver.To obtain different beam patterns, the backscatter tag is configured with specific -phase shifts between consecutive antennas and is rotated around its axis toward the receiver using a motor with fine angle control.The received signal power is then plotted as a function of the tag rotation angle.This process is repeated for various tag configurations, and the obtained results are compared with the theoretical beam pattern, as depicted in Fig.11(b).A maximum gain of 11 dB is observed, which is close to the theoretical gain of 12 dB [calculated as 10× log(16)] for a four-antenna array operating in backscattering mode.The slight deviation in the clock phases causes minor increases in side lobes within the measured beam patterns.With λ/2 antenna separation and 15 • of IF phase resolution, supported beam resolution is about 5 • , which is enough for demonstration purposes.However, the resolution requirement still depends on the range that certain applications need.Fig. 12(a) illustrates the setup for the outdoor wireless test.In this setup, a commercial off-the-shelf (COTS) WiFi transmitter is positioned 23-m away from the backscatter

FIGURE 11 .
FIGURE 11.(a) Beam pattern measurement setup and (b) measured WiFi beam pattern comparing with theoretical results.

FIGURE 12 .
FIGURE 12. (a) Wireless experiment setup, and (b) and (c) distance comparison between beam-steering approach and single-antenna approach.