Millimeter-Wave End-Fire Magneto-Electric Dipole Antenna and Arrays With Asymmetrical Substrate Integrated Coaxial Line Feed

A wideband vertically-polarized end-fire magneto-electric (ME) dipole antenna fed by an asymmetrical substrate integrated coaxial line (ASICL) is proposed for millimeter-wave (mm-wave) applications. The proposed ME dipole antenna has an impedance bandwidth of 60% and produces an end-fire radiation with a stable gain between 5.8 and 6.9 dBi in simulation. To verify the design concept and its application in arrays, a <inline-formula> <tex-math notation="LaTeX">$1 \times 8$ </tex-math></inline-formula> linear array with a fixed beam and a <inline-formula> <tex-math notation="LaTeX">$1 \times 4$ </tex-math></inline-formula> linear array with multiple beams are designed, fabricated and measured. The fabricated fixed-beam array prototype exhibits an impedance bandwidth of 62%, a peak gain of 15.9 dBi and a radiation efficiency of 79%. The multi-beam array is fed by a <inline-formula> <tex-math notation="LaTeX">$4 \times 4$ </tex-math></inline-formula> ASICL Butler matrix. Experimentally, excellent beam-scanning radiation patterns with reasonably low side lobes and low cross polarizations are demonstrated at 24, 30 and 36 GHz. And an overlapped operating bandwidth of 28.6% is achieved by the fabricated prototype.


I. INTRODUCTION
M ILLIMETER wave (mm-wave) technology is one of the most important parts of the emerging fifth generation (5G) wireless communications.Since the electromagnetic (EM) waves of mm-wave frequencies suffer from high propagation losses, high-gain antennas are usually required for mm-wave systems.Arraying is a typical and useful solution to enhance the antenna gain.In addition, to further improve the spatial coverage, beamforming or beam-scanning is another desirable property of antennas in mm-wave bands [1], [2].
Over the last two decades, many planner antenna arrays with broadside radiation have been reported for both highgain and beam-scanning requirements [3]- [10].But on the other hand, antenna arrays with end-fire radiation are still not common enough in mm-wave bands.End-fire antennas (arrays) can save the space and provide some flexibility in practical scenarios, and are attractive for various terminal devices.
Typical end-fire antennas reported in mm-wave bands include substrate integrated waveguide (SIW) horns [11], [12], Yagi-Uda antennas [13], [14], dielectric rod antennas [15], [16], and tapered slot antennas (TSAs) [17], [18].In [14] and [16]- [18], antenna arrays with a fixed beam were demonstrated, but these arrays were not suitable for beam-scanning applications.In [19], a circularly-polarized end-fire antenna, which was excited by an open-ended SIW, was presented with a beamscanning property.But the operating bandwidth was only 10%.A broadband printed dipole was reported in [20] and a wide-angle beam-scanning property was obtained by inserting a stub between adjacent elements to reduce the mutual coupling.However, the cross polarization was large and the feed network based on microstrip line (MSL) would suffer from high radiation losses in mm-wave bands [21], [22].In [23] and [24], flat lens antennas were investigated for high-gain and beam-scanning end-fire designs, but they were basically 3-dimensional (3D) structures and not easy for integration.Recently, the authors also reported a phased array exhibiting wide angle scan over a 3:1 bandwidth [25].However, a transverseelectromagnetic (TEM) horn was used and it was a little bulky and large in length.
By adopting the concept of magneto-electric (ME) dipole antenna, two end-fire SIW-fed antennas with vertical [26] and horizontal [27] polarizations respectively were reported.These two ME dipole antennas exhibited impedance bandwidths over 40%, but the multi-beam array designs were demonstrated with bandwidths narrowed to 20% due to employing the SIW feed networks.A dualpolarized antenna array combining those two ME dipoles in [26] and [27] was also reported by the authors in [28].The beam-scanning characteristic was examined through simulation, while the overlapped bandwidth was similarly limited to 18%.In [29], a substrate integrated coaxial line (SICL) fed ME dipole antenna was presented for endfire radiation.A wide impedance bandwidth of 50.7% was achieved by a 1 × 16 linear array and the beam-scanning characteristic was examined through simulation.However, the antenna elements were built in a vertically-oriented printed circuit board (PCB).Namely, each element was a broadside-type antenna and the whole array design had a 3D structure.More recently, another wideband end-fire ME dipole antenna was proposed and a 1 × 4 fixed-beam array was examined with an impedance bandwidth of 60.6% [30].However, this antenna was also fed by an MSL and the radiation was horizontally polarized.
In this article, a new wideband end-fire ME dipole antenna with vertical polarization is proposed for mm-wave applications.Compared to the previously reported end-fire ME dipoles, the proposed one exhibits a wider bandwidth and a smaller gain variation, and provides a simple interface with other planar circuits.The proposed ME dipole antenna contains a pair of shorted planar parallel plates serving as magnetic dipole and two pair of vertical metallized vias serving as electric dipole.Inspired by the earliest ME dipole antenna invented by Luk and Wong in [31], a -shaped probe is adopted to excite the antenna.To feed the proposed ME dipole antenna, an asymmetrical substrate integrated coaxial line (ASICL), which contains two dielectric substrates of different thicknesses, is co-designed as the feed line.In addition, a folded vertical wall consisting of metallized vias and strips is added to reduce the back radiation and to improve the gain of antenna.The feed structure containing the ASICL and the -shaped probe is similar to that in the authors' recent work in [25], but the antenna geometry, operating principle and antenna performance are remarkably different.To verify the design concept and its array applications, a 1 × 8 fixed-beam array and a 1 × 4 multi-beam array employing the presented ME dipole are designed, fabricated, and measured.Excellent characteristics are demonstrated by the fabricated prototypes.
This article is organized as follows.The characteristics of the ASICL are presented in Section II.The geometry and performance of the ASICL-fed end-fire ME dipole antenna are presented in Section III.The fixed-beam and multi-beam antenna arrays are demonstrated in Sections IV and V, respectively.Finally, the conclusion is presented in Section VI.

II. ASYMMETRIC SUBSTRATE INTEGRATED COAXIAL LINE
An asymmetric substrate integrated coaxial line (ASICL), which was firstly used by the authors in [25], is used as the feed line for the proposed antenna.Fig. 1 presents the configuration of an ASICL which consists of two dielectric substrates, top and bottom ground planes, a conductive signal line in the middle, and two rows of metallized vias placing at both sides of the signal line.Different to a conventional SICL, two Rogers 5880 substrates (ε r = 2.2, tanδ = 0.0009) with different thicknesses (h 1 = 0.254 mm, h 2 = 0.787 mm) are purposefully chosen in this study.In addition, a Rogers 4450F layer (ε r = 3.52, tanδ = 0.004, h b = 0.1 mm) is used as the bonding film in the middle of the two substrates.Due to the asymmetric geometry, the cross-section electric field distribution of an ASICL as depicted in the embedded diagram [Fig. 1, dash-line box] is similar to that of an MSL.As already studied in [25], the fundamental mode of an ASICL is the TEM mode and the first higher-order mode is TE 10 .In this study, the cut-off frequency of the first higher-order mode is 56.4 GHz, which is far away from the frequency range of interest.
A similar study of the characteristic impedance (CI) of the ASICL is carried out and more details are included in this article.It should be noted that the bonding layer (Rogers 4450F) is ignored during initial simulations since it has insignificant influence on the results.Fig. 2 presents the CI value of the proposed ASICL with different 'w in ' and 'w out ', which are the width of the center signal line and the distance between the two via walls, respectively.Some useful design guidelines are obtained from Fig. 2 as follows: (1) 'w out ' has little effect on the CI unless the via wall is very close to the signal line.Thereby, 'w out ' should not be too small for good tolerance.(2) When 'w out ' is within the range from 2 to 2.5 mm, the CI-w in curve keeps identical.With 'w in ' ranging from 0.1 to 1.4 mm, the CI varies from 25 to 108 ohms.(3) When 'w in ' is 0.6 mm, the CI equals to 50 ohms.One extra benefit of using unequal substrate thicknesses is that 'w in ' is not necessarily too narrow for achieving a relatively high CI.

III. ASICL-FED END-FIRE MAGNETOELECTRIC DIPOLE A. GEOMETRY AND OPERATING PRINCIPLE
The design procedure of the proposed ME dipole is illustrated in Fig. 3. Firstly, at the open end of an ASICL, the center signal line is extended out slightly and then connected to a -shaped probe through a metallized blind hole in the thinner substrate.The central parts of the upper and lower ground planes are extended with an identical length to form a shorted quarter-wave patch.This shorted quarter-wave patch can be excited by the -shaped probe and provide a horizontal magnetic dipole source at the open aperture.Secondly, two pairs of metallized vias are added at the two sides of the upper and lower ground planes [Fig.3 (b)].These metallized vias can also be excited by the -shaped and provide a vertical electric dipole source.Thirdly, two post walls are added to serve as a reflector.Finally, the reflector walls are effectively folded by adding the copper strips.
The -shaped probe as depicted in Fig. 3 (a) consists of a first strip portion on the lower surface, a second strip portion on the upper surface, and a through hole connecting the two strips.Specially, the first strip portion is grounded by the extended lower ground plane in the same layer and therefore performs as a coplanar waveguide (CPW) type transmission line.Since the majority of energy for an ASICL is distributed between the signal line and the closer ground plane, a good transition can be achieved between the ASICL feed and the -shaped probe.In this manner, the -shaped probe can easily carry the EM waves and excite the antenna.This also explains the reason why an ASICL instead of a conventional SICL is employed.
The detailed description of geometric parameters is presented in Fig. 4 and the corresponding dimensions are listed in the figure caption.As shown in Fig. 4 (a), the final design of the antenna consists of four Rogers 5880 substrates.The 1st and 2nd substrates support the ASICL and the -shaped probe.The 3rd and 4th substrates support the electric dipole and the folded reflector wall.The Rogers 4450F bonding layer is ignored in the simulation as explained before.The -shaped probe and the center line are highlighted in yellow for a friendly display.
For a better understanding of the operating principle of this antenna, the current distribution on the two pair of metallized vias and the electric field distribution at the antenna slot aperture are presented in Fig. 5.It should be noted that the presented antenna is analyzed separately with the radiation boundary at all outer sides.At the moment of t = 0, strong dipole-like currents are excited on the metallized vias.At the same time, a strong electric field is also excited at the antenna aperture, which is equivalent to a horizontal magnetic current.At the moment of t = T/4, both the currents on the vias and the electric fields at the slot aperture get weak.Therefore, a pair of orthogonal electric dipole and magnetic dipole are excited simultaneously.Namely, an ME dipole is excited as expected.

B. PERFORMANCE OF THE ELEMENT ANTENNA
The simulated reflection coefficient (S 11 ) and gain of the proposed ME dipole are presented by Fig. 6.The simulated impedance bandwidth is 59.4%  A comparison between the proposed and previously reported end-fire ME dipole antennas is carried out in Table 1.All the listed antennas are designed for mm-wave frequencies and they are basically based on different feed techniques.Note that all the listed values in Table 1 are simulated results.The impedance bandwidth of this work is comparable with that of [30], and is larger than those of designs in [26] and [27].The peak gains of different antennas are similar with each other, but this work exhibits a much smaller gain variation of 1.1 dB.Due to the unbalanced characteristic of the ASICL, the cross polarization of this work is not such small as those of designs in [26] and [27] based on the SIW, but it is much smaller than that of the design in [30] based on the MSL, reaching a reasonably low level of −19 dB.Furthermore, the front-to-back ratio (FBR) is comparable with reported ones.

C. PARAMETER STUDY AND EFFECT OF THE FOLDED REFLECTOR WALL
To provide useful design guidelines for the proposed antenna, some important parameters are studied.It should be noted that the folded reflector wall is not included in the beginning, i.e., the study model is as depicted in Fig. 3 (b).
The extended part of the parallel ground planes operates as a shorted quarter-wave patch that determines the magnetic dipole source.Fig. 8 presents the simulated reflection coefficient of the antenna with different extension lengths 'l a '.The upper resonant frequency decreases while the lower one keeps unchanged with increasing 'l a '.Thus, a conclusion can be made that the upper resonant frequency is due to the magnetic dipole.
The width of the extended parallel planes 'w a ' determines the width of the quarter-wave patch.This parameter also   affects the resonance of the magnetic dipole.Fig. 9 presents the simulated reflection coefficient with different 'w a '.It turns out that the upper resonant frequency decreases with increasing 'w a '.Fig. 10 presents the simulated reflection coefficient with different thicknesses of the 3rd (and 4th) substrate 'h 3 ', which determines the length of the electric dipole.The lower resonant frequency decreases with increasing 'h 3 '.At the same time, the upper resonant frequency remains nearly unmoved.It is also shown that the lower resonant frequency is determined by resonance of the electric dipole.Substrate tickness of h 3 = 1.575 mm is selected because of the commercial availability.
Extending the substrate is an effective method to improve the impedance matching and enhance the gain.Fig. 11 presents the simulated reflection coefficient and  gain with different lengths of the substrate extension 'l e '.By choosing l e = 2 mm, both the impedance matching and antenna gain can be improved remarkably.On the other hand, the two resonant frequencies show insignificant shifts.
Finally, the effect of the folded reflector wall is studied.Fig. 12 presents the reflection coefficients, FBRs and gains of three different cases: no reflector wall, with the post wall and with the folded wall.For the reflection coefficient [Fig.12 (a)], three curves are almost same.While for the FBR [Fig.12 (b)], by adding the post wall, the FBR is improved significantly over a wide frequency band.Then, by folding the wall with a suitable length, the FBR at the lower frequency band is further improved.Furthermore, owning to the reflector, the gain [Fig.12 (c)] is enhanced remarkably.And by properly folding the walls, the gain is also stabilized over the operating band.

IV. FIXED BEAM ARRAY A. CONFIGURATION
A 1 × 8 linear array with a fixed beam is examined in this section.Fig. 13 presents a top view of the array antenna.The element space is 6 mm, 0.6λ 0 of the center frequency of 30 GHz.With this element space, the mutual coupling between different elements is weak and has little effect on the array performance.A 1-to-8 ASICL power divider is designed as the feed network by cascading 1-to-2 dividers.At the input end, an ASICL-to-MSL transition is introduced for the power input via the commercial end-launch connector during the test.Several holes are located at two sides to fix the multiple PCBs.

B. FEED NETWORK
For the 1-to-8 ASICL divider, the design procedure is similar to a MSL-type divider.The key point is to determine the widths of center signal lines for good impedance matching.In this study, the CI-w in curve in Fig. 2 provides a useful   the simulated directivity, an average radiation efficiency of 79% is obtained.
Fig. 17 presents the normalized radiation patterns of the array at 23, 30 and 37 GHz.For co-polarization patterns [Fig.17

D. COMPARISON
A comparison between the proposed and reported mm-wave end-fire antenna arrays is carried out in Table 2. Owning to the wideband element and ASICL-based feed network, the  proposed array exhibits the largest impedance bandwidth exceeding 60%.Furthermore, the peak gain of this design is larger than those of designs in [17] and [18].Moreover, by using the ASICL feed technique, the proposed array obtains a high radiation efficiency of 79%, which is comparable to that of the SIW-based design in [18].

V. MULTI-BEAM ARRAY A. CONFIGURATION
To verify the beamforming potentials of the proposed ME dipole antenna in phased arrays, a multi-beam array is designed and examined in this section.Fig. 18 presents the configuration of the multi-beam antenna array.A 1 × 4 linear array is fed by a 4 × 4 ASICL Butler matrix (BM).The element space is determined to be 5.4 mm, 0.54λ 0 of 30 GHz, for reasonably low mutual coupling.In addition, a dummy element is added at each side in order to reduce the influence of edge effect.The dummy port is left to be opened since extremely low energy arrives at it.The substrate was grooved to reduce the influence of surface waves during beam scanning, while it was found later that this groove structure did not eliminate the surface wave but shifted the suspect frequency and enlarged the operating bandwidth slightly.This will be briefly discussed in the end of this section.

B. 4 × 4 BUTLER MATRIX
As shown in Fig. 18, the 4 × 4 BM consists of four 3-dB hybrid couplers, two crossovers, two −45 • phase shifters, and two 0 • phase shifters.All of these phase shifters are in terms of the phase delay introduced by the crossover.Each component for this BM is carefully designed with a wideband operation.The detailed design guidelines of the BM are not included in this article and interested readers can refer to literature [32] for relative studies.Fig. 19 (a) and (b) present the simulated amplitudes and phases of the S-parameters of the 4 × 4 BM.Overall, reasonably good amplitude and phase responses are achieved over a wide frequency band, although the at near 33 GHz are not such perfect.At the lower (24 GHz), center (30 GHz) and upper (36 GHz) frequencies, the worst amplitude unbalance is 2.2 dB and the biggest phase error is 11 degrees.

C. MEASUREMENT AND PERFORMANCES
Fig. 20 shows the fabricated prototype of the multi-beam antenna array.The fabrication process is the same as last section.The S-parameters and radiation performances of the multi-beam array are measured.During each measurement, those untested ports are terminated by 50-ohm loads.Normalized radiation patterns at 24 GHz, 30 GHz and 36 GHz are presented in Fig. 22, where solid lines and dash lines represent the simulated and measured results respectively.The measured and simulated radiation patterns are in good agreement.The array steers the main beam at different azimuth angles when different ports are excited separately.The worst side lobe level is −6 dB and the cross polarization maintains below −20 dB.The scan angles and gains are summarized in Table 3. Due to the geometric symmetry, only results when exciting Port #1 and Port #3 are listed.At 24 GHz, the array obtains the largest scan angles, ±19 • and ±51 • .With the frequency increasing, the scan angle reduces.The measured and simulated gains listed in Table 3 also agree with each other reasonably well.At all of these three frequencies, the gain variation due to beam-scanning is smaller than 0.9 dB.

D. DISCUSSION AND COMPARISON
The simulated gain curves versus frequency are plotted in Fig. 23.It can be observed that there exists a substantial     dropoff near 34.5 GHz when Port #1 is excited and near 32.75 GHz when Port #3 is excited.This dropoff is caused by the grating lobe condition [33], [34] where a significant surface wave mode is generated along the arraying direction.The groove structure mentioned above does not eliminate this surface wave effectively but delays its happening.A possible solution of avoiding the grating lobe issue is to reduce the element space but the mutual coupling should be considered carefully.This issue is not discussed in detail in this article and we will try to address it in future works.
A comparison between the proposed and reported endfire multi-beam arrays is carried out in Table 4.All of the listed arrays contain a Butler matrix as the feed network.Considering the gain dropoff issue, the operating band of this array can be defined as 24−32 GHz.The relative bandwidth is 28.6%, which is wider than those of the designs in [26] and [27].This work also achieves a smaller gain variation and a comparable scan range with that of the design in [26].The SLL is medium and reasonably low.

VI. CONCLUSION
An ASICL-fed end-fire ME dipole antenna with wide bandwidth and superior radiation characteristics has been proposed and studied.The antenna is constructed in stacked laminates and can be fabricated using the lowcost PCB process.Two antenna arrays employing this proposed antenna are examined through experiments, including a fixed-beam array and a multi-beam array.The fabricated fixed-beam array prototype obtains a wide bandwidth of 62%, a large peak gain of 15.9 dBi, and a high radiation efficiency of 79%.The multi-beam array produces four separated beams by adopting a 4 × 4 Butler matrix.The fabricated prototype demonstrates excellent beam-scanning characteristics over a 28.6% bandwidth.The cross polarization and the side-lobe level are reasonably low.With these advantages, the proposed antenna and arrays could be attractive candidates for many mm-wave applications, including the 5G and beyond.

FIGURE 1 .
FIGURE 1. Perspective view of an ASICL with two substrates of different thicknesses and the electric field distribution at the cross-section [in dash-line box].Dimensions: d = 0.4, s = 0.8, in mm.

FIGURE 2 .FIGURE 3 .
FIGURE 2. Characteristic impedance of the ASICL with different 'w out ' and 'w in '.

FIGURE 5 .
FIGURE 5. Current and Electric-field distribution of the proposed antenna at t = 0 and t = T/4, where T is the time period.The antenna is analyzed with the radiation boundary at all outer sides.

FIGURE 6 .
FIGURE 6. Reflection coefficient and gain of the proposed ME dipole antenna.
(21.3 to 39.3 GHz) with |S 11 |< − 10 dB.Within this operating band, the antenna gain varies from 5.8 to 6.9 dBi, with a variation of 1.1 dB.Normalized radiation patterns at 30 GHz are illustrated in Fig. 7.The Co-polarization patterns in E-plane and H-plane are almost identical with a halfpower beamwidth (HPBW) of 93.1 • and 92.5 • , respectively.The cross-polarization and back radiation levels are below −19.2 dB and −20.4 dB, respectively.

FIGURE 7 .
FIGURE 7. Normalized patterns of the proposed ME dipole antenna at 30 GHz.

FIGURE 8 .
FIGURE 8. Simulated reflection coefficient with different patch lengths 'l a '.

FIGURE 9 .
FIGURE 9. Simulated reflection coefficient with different patch widths 'w a '.

FIGURE 11 .
FIGURE 11.Simulated reflection coefficient and gain with different lengths of substrate extension 'l e '.

FIGURE 12 .
FIGURE 12. Simulated reflection coefficient, FBR and gain of three different cases.

FIGURE 16 .
FIGURE 16.Measured and simulated reflection coefficients and gains, and the simulated directivity, of the 1 × 8 linear array.
, left column], good agreements are achieved between the measurement and the simulation at different frequencies.A narrow beam is obtained in H-plane because of the linear array arrangement.The measured side lobe level keeps below −13 dB.For cross-polarization patterns [Fig.17 , right column], the measured cross polarizations are below −25 dB at 23 GHz and 30 GHz, and below −20 dB at 37 GHz.These measured results are not such small as simulated ones probably due to the dynamic range limitation of the test system.

FIGURE 18 .
FIGURE 18. Configuration of the multi-beam antenna array.

Fig. 21
Fig. 20 shows the fabricated prototype of the multi-beam antenna array.The fabrication process is the same as last section.The S-parameters and radiation performances of the multi-beam array are measured.During each measurement, those untested ports are terminated by 50-ohm loads.Fig. 21 presented the measured S-parameters of the multibeam array.Due to the geometric symmetry, |S 11 | and |S 44 | equal with each other approximately, and so do |S 22 | and