Wideband and Dual-Polarized Crossed Dipole Antenna Design Using Substrate-Integrated Coax

A wideband and dual-polarized dipole antenna design technique is presented. The proposed antenna employs two sets of dipole elements in orthogonal configuration and separated by a dielectric layer. The dipole elements are excited with two separate substrate-integrated-coaxial feeds to achieve high port-to-port isolation and the overall antenna stackup is enclosed in a substrate-integrated-waveguide (SIW) cavity. To validate the proposed substrate-integrated-coaxial fed dual-polarized antenna design method, a prototype antenna was fabricated using standard multi-layer printed circuit board (PCB) manufacturing technologies to operate in the Ku-band. Measured and simulated results exhibit that the designed antenna has a wide impedance bandwidth of less than -10dB reflection coefficients at 10.28-14.8GHz and the port-to-port isolation between the two orthogonally polarized dipole antenna ports is greater than 20dB at most frequencies, making it suitable for many applications such as satellite communications. The antenna has a relatively stable gain pattern with a gain value of 5.5-6.5dBi and a low cross-polarization level (lower than 16dB) over the operating frequency range.


I. INTRODUCTION
W ITH the rapid advances in modern wireless communication systems, the increasing demand for high data rate is becoming a critical issue in many applications such as mobile communications and satellite communications system. The wideband and dual-polarized antennas have received significant attention because of its ability to offer polarization diversity and are being adopted in many multifunctional RF antenna systems to significantly enhance the channel capacity and spectral efficiency, reduce the multipath fading, and maintain reliable wireless communication links.
When designing standalone dual-polarized antenna element for real-world antenna applications, there are multiple electrical and mechanical challenges: 1) the main electrical challenge includes achieving wider impedance bandwidth with stable gain patterns over the operating frequency range, maintaining high orthogonal port isolation and low cross-polarization level to maintain the polarization purity. 2) Besides these electrical challenges, there are also many mechanical and manufacturing challenges such as keeping the antenna profile low with compact size and light weight, ability to manufacture the antennas in large number by leveraging the existing low-cost manufacturing technologies such as the printed circuit board (PCB) manufacturing, and easier integration of the antenna elements with external RF electronics hardware which are extremely important to successfully employ these dual-polarized antenna elements for real-world antenna applications.
In this paper, a new way of designing and implementing a standalone dual-polarized and wideband dipole antenna element based on substate-integrated coaxial feed technique is proposed and investigated. To demonstrate the proposed dual-polarized antenna design concept, an antenna structure comprising two sets of planar dipole element in orthogonal configuration and excited by two separate substrate-integrated coaxial feeds is designed and manufactured. By utilizing the substrate-integrated coaxial feed technique, high port-to-port isolation (>20dB) and wider impedance bandwidth (approximately 36%) are achieved. The proposed antenna design method is simple and can easily be manufactured in large quantities by leveraging the existing low-cost printed circuit board (PCB) manufacturing technologies, and also, has easier assembly and integration with the backend RF electronics hardware. This paper is divided into three sections. In Section II, the proposed antenna design method and simulated performance is shown. Section III presents the measurement of a prototype antenna, and a conclusion is drawn in Section IV.

A. ANTENNA CONFIGURATION
The proposed antenna configuration is shown in Fig. 1(a). It is a multi-layer structure and has two sets of planar   modified bowtie-shaped crossed dipole elements printed on the top surfaces of two separate dielectric layers separated by a prepreg bonding layer ( Fig. 1(b-c)). To generate orthogonal polarization and maintain high cross-polarization level, the dipole elements are configured in a perpendicular orientation in which the two feed sources exciting the two dipole elements are located in close proximity ( Fig. 1(a-b)) resulting in strong port-to-port coupling. To mitigate the port-to-port coupling and improve the orthogonal port isolation, the dipole pairs are excited with two separate substrate-integrated-coaxial feeds ( Fig. 1(b-c)). Three plated via holes with a radial via-to-via distance of 70.57mil were embedded in the substrate material to realize the substate integrated coax. Each substrate integrated coax has a radius of 40.57mil which corresponds to a characteristics impedance of 52 .
The overall antenna structure is enclosed in a square shaped substrate-integrated waveguide (SIW) cavity to enhance the antenna gain value and minimize the backward radiation. In the SIW cavity, four periodic rows of plated via fence are embedded in the dielectric material to create a square cavity. These plated via rows along with the top and bottom metallic layers ( Fig. 1(c)) introduce a structure like a square metallic cavity. Fig. 1(b-c) and Fig. 2 shows the detailed configuration of the dipole elements and the antenna stackup. The stackup is made up of seven 'TSMDS3B' (dielectric constant ε r = 2.95, loss tangent δ = 0.0011) dielectric materials of different thicknesses (Fig. 1(c) & Fig. 2) with six 4milthick 'RO4550' (dielectric constant ε r = 3.52, loss tangent δ = 0.004) prepreg bonding layers and seven 0.5oz copper layers. To prevent possible electrical shorting during the electroplating and lamination process, several non-functional via pads were implemented on the anti-pad layers ( Fig. 1(c)). The two dipole elements of the antenna structure ( Fig. 1) are excited through two sub miniature push-on micro (SMPM) RF connectors. The two feed vias of the proposed dual polarized antenna structure are located in close proximity which complicates the connector assembly due to narrower spacing between the two feed vias of the antenna structure. To facilitate the connector integration, a stripline layer with two extended transmission lines is implemented underneath the antenna ground plane. The stripline layer includes a via fence (Fig. 1(c) & Fig. 2) between the two-transmission lines to mitigate the interference between the two orthogonally polarized electric field lines. By adjusting the length of the two transmission lines, the relative spacing between the two antenna ports can be made flexible to accommodate the RF connectors and ease the external RF electronics hardware integration.

B. SIMULATION RESULTS
The proposed antenna structure was designed and simulated using Ansys' electromagnetic solver High Frequency Structural Simulator (HFSS) version 2022R1 software package. All the design parameters including the via diameters, minimum via-to-via distance (10mil) and via pads were optimized to meet the printed circuit board (PCB) manufacturer's design guidelines and Table 1 presents the dimensions of the different parameters of the designed antenna structure. Fig. 3a demonstrates the simulated active reflection coefficients (active S ii = S ii + i =j S ij ) and the port-to-port isolation (|S 21 |) between the two orthogonally polarized antenna ports. Fig. 3b shows the Smith chart of simulated S 11 and S 22 . From the simulated predictions, it is obvious that the designed antenna structure has a relatively wide impedance band-width with the |S 11 | and |S 22 | less than −10dB over 10.28-14.78GHz frequency band and the port-to-port coupling (|S 21 |) is less than −20dB across most of the frequency band. The fractional impedance bandwidth is about 36%.
To predict the radiation patterns and calculate the crosspolarization, the designed antenna structure was simulated in both dual-linear (Vertical/Horizontal) polarization and dualcircular (Right-hand/Left-hand) polarization configurations, and the antenna gain patterns in both configurations were predicted in HFSS. For the configuration of dual linear polarization, one antenna port (vertical or horizontal) was excited at a time and the other port was terminated with a 50 resistive load. For dual circular-polarization configuration, both the antenna ports were excited simultaneously with a 90 • phase difference between the two orthogonal ports. Fig. 4(a-d) shows the simulated two-dimensional antenna radiation patterns along the XZ and YZ planes at 10.7GHz and 13.75GHz for dual linear polarization configuration and Fig. 4(e-f) shows the radiation patterns for dual circular polarization configuration at 10.7GHz and 13.75GHz. The simulated radiation pattens at 12.5GHz is shown in Fig. 5(a) as a function of the elevation angle and Fig. 5(b) presents the co-pol and cross-pol broadside peak gain patterns as a function of frequency. From the simulated predictions, it is observed that the designed antenna has a 16dB lower cross-pol. gain level over most of the frequency band.

C. PHASE CENTER OF THE DUAL-POLARIZED DIPOLE ELEMENTS
The two orthogonally polarized dipole elements (i.e., the vertical and horizontal dipoles) in the proposed antenna structure are positioned slightly offset (by 52.9mil) relative to the center of the antenna structure to accommodate the substrate integrated coaxial via fences and this results in a phase center offset between the horizontal and vertically polarized dipole elements. For the proposed dual-polarized antenna structure to be useful for practical applications, it is necessary to evaluate the effect of the phase center offset on the antenna polarization purity over the operating frequency range. The phase center of an antenna is a point at which the far field emanates, i.e., all the electromagnetic field waves radiated from the antenna elements have equal phase front along the spherical curvature at that imaginary point [37], [38], [39].
To calculate the phase center variations of the designed antenna structure using HFSS software, the radiation boundary of the designed antenna structure was moved in along the z-axis and the phase variation over the beamwidth was minimized using HFSS. Fig. 6(a) shows the simulated phase center variations of the two linearly polarized antenna element as a function of the scan angle at 11GHz. As seen, the phase offset between the H-pol. and V-pol. dipoles is relatively smaller (9 • ) along the boresight and has little effect on the axial ratio performance at the operating frequency range. Fig. 6(b) presents the simulated phase center variation over the frequency range. As evident, the phase center variation of the two orthogonal dipole elements is relatively stable over the entire operating frequency range. Authorized licensed use limited to the terms of the applicable license agreement with IEEE. Restrictions apply.

III. MEASUREMENT AND VALIDATION
A prototype of the designed multi-layer antenna structure was manufactured using standard printed circuit board (PCB) manufacturing technique and assembled with two surface mounted standard sub miniature push-on micro (SMPM) connectors at the two excitation ports. Fig. 7 shows a picture of the fabricated antenna prototype. A 4-port vector network analyzer (Rhode and Schwarz's ZNB40) was used for measuring the reflection/transmission coefficients and the far-field radiation patterns. The measured antenna gain of the fabricated antenna, in both linear/circular polarization configuration, was calculated by comparing with a Ku-band standard gain horn antenna. To calculate the circularly polarized gain patterns, a COTS 90 • hybrid coupler was used to generate the 90 • phase difference between the two orthogonal (vertical/horizontal) antenna ports.

A. MEASURED RESULTS IN DUAL LINEAR POLARIZATION CONFIGURATIONS
The measured results of the reflection coefficients and portto-port isolation are shown in Fig. 8 (a-b) and compared with the simulated predictions. From the comparison, it is seen that the measurements agree well with the HFSS simulations. The simulated impedance bandwidth for |S ii | lower than −10dB is 36% (10.28-14.8GHz) while the measured |S ii | is less than −10dB from 9.9GHz to 14.5GHz, corresponding to a fractional bandwidth of 37.7%. The measured port-toport isolation is greater than 20dB at most frequencies over 9.7-14.7GHz frequency range.
To measure the radiation patterns in dual linear polarization configuration, the vertical and horizontally polarized antenna elements were excited at a time while the other one was terminated with a 50 load. To calculate the gain patterns as a function of angle and frequency, the testing antenna was rotated under computer control and on the receive end, a standard gain horn antenna was used and oriented in vertical and horizontal configuration to calculate the co-pol/crosspol gain patterns of the testing antenna. Radar absorbing materials was placed all around to reduce any wanted reflections. The measured and simulated co-pol/cross-pol gain patterns of the fabricated antenna at 12.5GHz and 13.5GHz are shown in Fig. 9 for vertical/horizontal polarization and Fig. 10 presents the gain patterns as a function of frequency. As seen, the gain patterns are relatively stable and symmetric around the broadside peak gain, and the cross-polarization levels are approximately 16dB lower than the broadside peak gains. The smaller deviations and ripples in the measured results can be attributed to the reflections and losses in the RF cables, connectors, attenuators, and the fabrication tolerances.

B. MEASURED RESULTS IN DUAL CIRCULAR POLARIZATION CONFIGURATIONS
To generate right-hand and left-hand circular polarization (RHCP and LHCP) from the proposed dual linearly polarized antenna structure, a COTS 90 • hybrid coupler was connected to the two linearly polarized antenna ports (i.e., H/V ports) to generate a 90 • phase difference between the two orthogonal dipole elements as shown in Fig. 11(a) [40], and the axial ratio performance and RHCP/LHCP gain patterns were measured. Fig. 11(b-c) compares the measured and simulated co-pol/cross-pol (i.e., RHCP/LHCP) gain patterns and the corresponding axial ratio as a function of frequency. The designed antenna with the 90 • hybrid coupler exhibits a wide axial ratio bandwidth and the 3dB axial ratio bandwidth is about 34.5%. Table 2 compares the performance characteristics of the proposed dual-polarized crossed-dipole antenna structure with other PCB-based dual-polarized antennas. It is 370 VOLUME 4, 2023 evident that the proposed standalone dual-linearly polarized antenna achieves relatively wide impedance bandwidth with low cross-polarization level and high port-to-port isolation.

IV. CONCLUSION
In this paper, a new way of designing and implementing a wideband and dual-polarized standalone antenna element has been demonstrated. The antenna design method is straightforward and employs two sets of substrate-integrated coaxial feeds to improve the orthogonal port isolation between the two antenna ports. Measured results show that the proposed antenna has a relatively wider impedance bandwidth (about 36%), high orthogonal port isolation (>20dB), and low cross polarization level (lower than 16dB). The radiation patterns are relatively stable and symmetric over the operating frequency range. The proposed antenna structure is mechanically robust and has planar configuration which makes it easily integrable with the external RF electronics hardware without having any integration issues and cable requirement. Also, the proposed multi-layer antenna structure can easily be manufactured in high-volume by leveraging existing low-cost PCB manufacturing technologies.