A Novel and High-Gain Switched-Capacitor and Switched-Inductor-based DC/DC Boost Converter with Low Input Current Ripple and Mitigated Voltage Stresses

High voltage gain DC-DC boost converters are used widely in grid-connected applications through integration with the Renewable Energy Sources (RESs). Photovoltaic (PV) arrays or Fuel Cells (FCs) generate a limited value of the DC voltages and then for high power and high voltage applications, at the first stage, these voltages should be increased. This study presents a high gain, single-switched, and efficient DC-DC boost converter using the switched-capacitor and switched-inductor cells. These blocks easily can enhance the voltage and present an input current with the less values of the ripples. This will be done through replacing the location of the input inductors and by applying a switched-inductor block. Magnetizing in parallel and demagnetizing in series for the inductors present the smaller input current stresses. A single switch is used for the proposed boost converter that directly decreases the complexity of the control circuit for obtaining a fixed DC voltage at the output side for flexible input voltages or loads. More voltages will be presented by the used switched-capacitor cell simply by adding several diodes and capacitors. A deep and detailed mathematical analysis will be presented for continuous (CCM) and discontinuous conduction modes (DCM) and a 200 W laboratory-scaled prototype is presented. The results of the hardware tests confirm the correctness of the theoretical analysis and simulation results.


I. INTRODUCTION
High voltage gain DC-DC converters widely are used in industrial, grid-connection, and Renewable-based applications [1][2][3]. Since some kinds of the RESs like the Photovoltaic (PV) arrays and Fuel Cells (FCs) can only generate limited values of the DC voltages, for any kind of grid or industrial applications, these voltages should be increased through a power converter circuit. For that, many types of step-up converters have been presented [4][5][6]. Conventional DC-DC boost converters are including an inductor as the magnetic field storage, a capacitor as the electric field storage, and a power switch and diode as the semiconductor components. This structure can double the input voltage for 50 percent of the duty cycle of the power switch. Although, the higher voltages can be obtained for longer duty ratios, for many of the applications, this voltage is not enough and leads to high power losses. For example, for a PV panel with 24 VDC, by a conventional DC-DC boost converter, a voltage closed to 48 VDC can be obtained. Furthermore, the power values that can be transferred through a classic converter are less than the required value for high-power applications. For that, many different types of DC-DC boost converters have been presented. Isolated or non-isolated boost converters, unidirectional and bidirectional converters, voltage or current-fed structures, converters with hard or soft Pulse Width Modulation (PWM) switching techniques, and minimum and non-minimumphase converters are among the presented topologies. All of these converters have advantages and disadvantages. For example, the non-isolated converters normally use a transformer to enhance the voltage [8][9]. For that, at the first stage, an inverter block is applied to convert the input DC voltage to an AC one, and at the second stage, this voltage is increased by the transformer. After that, a rectifier block is used to converter the generated high amplitude AC voltage to the desired DC voltage. As can be predicted, different inverter, transformer, and rectifier blocks should be used that directly enhance the number of the used components, cost, weight, and power losses. Unidirectional converters are including both the power switches and diodes and only can convert the DC to the AC or AC to the DC voltages [10][11]. As can be found, these types of converters are not suitable for many of the onboard applications such the vehicle to grid (V2G) or grid to vehicle (G2V) utilizations as a part of the electrical vehicles (EVs) applications [12][13][14][15][16]. Instead, in the unidirectional boost converters, diodes are replaced by the power switches and a control algorithm and a circuit should be added to the converter to activate or deactivate the replaced switches when should be in ON or OFF-states. These converters are suitable for many of onboard applications like the sensor applications or EVs, but a control prototype circuit should be designed and applied. Voltage and current-fed types of the converters normally can decrease and increase the input voltage at the output side respectively [17][18][19]. This means the voltage-fed converter acts as a Buck converter and includes an input capacitor whereas the current-fed converter includes an inductor at the input side and enhances the input voltage. These converters normally use a transformer between the input and output blocks that means they need the inverter and rectifier blocks that easily can present higher power losses and lower efficiency values. The main part of the power losses is the switching one. The dynamic losses can be presented and calculated through a current with a power of two through the internal resistance of the components. This type of power loss can be presented for all passive and active semiconductor devices like the capacitors, inductors, switches, and diodes. But in comparison with the switching losses, the dynamic losses are completely smaller. This comes back to the cross points in voltage and current curves of the semiconductors. It means for the hard switching techniques, the power losses for the time intervals that a switch or diode is turning on or off completely has a high loss value. For ideal switching conditions, when a switch is activated, the current enhances and voltage decreases through a step waveform. But in practice, it can be seen the time intervals that both of the current and voltage waveforms exist. By increasing the switching frequency, the number of the intersection time intervals will be increased and gives more switching power losses. For that, different soft switching techniques like the zero voltage (ZVS) and zero current switching (ZCS), and zero voltage (ZVT) and zero current transition (ZCT) are presented [20][21][22][23]. Although these techniques help the switches and diodes to work under fewer power losses, normally the design of these snubber structures is difficult and the switching frequency of the converter extremely is high. Also, more components should be added to the converter. Furthermore, the voltage gain is an important issue that should be considered. Charge pump or switched-capacitor (SC) circuits simply can be added to the converter to enhance the gain of the converter. In this technique, the energy transferring issue is done by the capacitor-diode (C-D) or capacitor-switch (C-S) blocks and no more inductors are added. The most important advantages of the SC blocks are the monolithic integration capability and enhancing the voltage through the simple (C-D) or (C-S) cells. In fact, the application of more SC blocks can present a higher DC voltage. To simplify the SC-cell configuration, one technique is using the diodes instead of the transistors. Although the dynamic loss of the diode normally is more than for a transistor, the control of the transistors can need a difficult algorithm and implementation. Also, by using the fast recovery silicon carbide (SiC) and Gallium Nitrite (GaN) types of the power diodes, this loss can be decreased to a large extent. Different studies are investigating the SC configurations. [24] presents a configuration with six power switches, three floated and one grounded capacitors, and totally four power diodes. The most advantage of this converter can be summarized as receiving to the low-ripple input currents and variable voltage ratio based on the applications. [25] presents an SC cell-based converter with totally two power switches, three floating and three grounded capacitors, and six power diodes. The soft switching capability of the proposed converter and less current stresses for the proposed SC block is among the most important features of the proposed configuration. The converter in [26] has a total of ten power switches, three floated, and one grounded capacitor without any power diodes. Bypassing the fault in onboard form, and the possibility to be connected to multiple loads are the important specifications of the converter. The converter in [27] presents a high voltage gain and lower power losses based on the small current stresses for the power components through the eight power switches, and four floated capacitors. The SC configuration will be activated through the two different switching approaches. Simple control schematic, including only two power switches and a variable and flexible voltage gain property, are among the most important advantages of the converter in [28]. This converter uses six grounded capacitors and six power diodes. Another technique to enhance the voltage by a boost converter is using the switched-inductor (SL) blocks. By this technique, normally a pair of the same inductors are used and magnetized in parallel and demagnetized in series. By this method, the specified capacitor will be charged and then a higher output voltage is obtained at the next working steps. A single core can be used for both inductors to minimize the weight and cost of the converter. This is an appropriate aspect for the proposed configuration since the weight and cost are among the serious problems of the inductors. Different SL-based circuits can be found in the literature. . The recent equation shows that the gain of the converter can enhance from two to three at the = 0.5 through the basic SC-cell. This rate can be enhanced by the elementary-lift cell converter to   [29]. In order to take advantage of the and −based DC-DC power boost converter, this study presents an SC-S -based power DC-DC boost converter with a high voltage gain including only on power switch. This converter can present a voltage at the load side equal to 7.5 t greater than the input voltage for c = 0.5. More voltage gains can be obtained through the longer duty ratios or more SC cells. For example, for the = 0.8, approximately, twenty times the input voltage can be reached at the output, or for = 0.5, by two cascaded SC-cells, more than twelve times the input voltage can be reached. The enhancing of the voltage is done by a simple diode connection configuration that eliminates the need for the supernumerary power switches. A PI-based controller technique is presented to fix the DC voltage under the variable input voltages or variable loads working conditions. A comprehensive theoretical investigation, simulation, and hardware test results are presented and confirm the reliability of the proposed converter for different working situations. Figure 1a illustrates the fundamental configuration for the SL-cell. As can be seen, a pair of the inductors are in connection with three diodes to enhance the voltage. For that, normally a pair of the same inductors are considered to be magnetized in parallel and demagnetized in the series . This means ℎ the diodes D1 and D2 are activating simultaneously and for the time intervals that the diode D3 is deactivated. The state of the diodes for parallel and serial charging and discharging operational modes are seen in figures 1b and 1c respectively. For minimizing the volume of the inductors, since the same inductors are used normally, a common core is applied. This can decrease the weight and cost of the configuration. Application of a pair of the inductors can decease the current ripples for the input source that is an important feature in DC-DC converters. Figure 2a presents a conventional SC-cell. As can be seen, when the switch S1 is activated the voltage across the capacitor C1 is equal to the input voltage . At the next step, this voltage is transferred across the capacitor C2 by the switch S2. After several time intervals, this voltage completely can be transferred across the load. Also, there are different serial SC blocks that can be used in multiphase approaches to enhance the input voltage. Figure 2b presents a multi-phase SC-cell. For this figure, when the switches S1 and S2 are in phase 1, the input voltage is transferred across the capacitor C1, and then by changing the location of the phases for the switches, a voltage equal to twice the input voltage is transferred across the second capacitor for the load. The higher DC voltages can be obtained through a cascaded SC-cell configuration. This state can be seen in figure 1c. This figure shows that when the switches are in mode I, the voltage across the capacitor C1 is equal to . For the next working phase and by changing the states of the switches to phase II, capacitor C1 is connected in series with the input voltage , therefore a double voltage of the is transferred to the capacitor C2. This state will be continued for the next time intervals and finally a voltage with fourth times greater than the input voltage appears across the capacitor Co. for the proposed converter in this study, a novel SC-cell is presented that can double the input voltage.

A. Proposed SL and SC-cell-based boost converter:
Firstly, in figure 3, a flowchart is presenting to show the different steps of this study. As can be seen, after presenting the proposed topology, the theoretical analysis is done for both CCM and DCM working modes, and then the simulation and hardware test results are compared with these results. Figure 4a shows the proposed converter. The proposed DC-DC boost converter as can be seen is formed by using a power switch (S1), five power diodes (D1, D2, D3, D4, and D7), two inductors (L1, and L2) and two capacitors (C1, and Co). For this side, the location of the switch, inductors and diode D1 is changed. Also, a switchedinductor block has been added to enhance the gain of the boost converter. The SC-cell is formed from two capacitors (C2, and C3) and two diodes (D5, and D6). This part is enhancing the voltage through the power diodes instead of the switches that can decrease the complexity of the control circuit and is among the most important specifications of the proposed SC-cell. The reaction of this part of the proposed converter is analyzed in this section through two different operating modes of the power switch. This can be followed through figures 4b and 4c. Figures 4b and 4c show the state of the converter when the switch is activated and deactivated respectively.

B. Gain calculation for the converter:
For time interval that the switch is activated, both inductors L1 and L2 start to be magnetized through the switch, and diodes D2 and D3. For this times, capacitor C1 is discharged since the current is established and passes from the negative pole of this capacitor and capacitor C2 is charged through the power diode D5 to be ready for discharging across the load for the next time interval. Also, the diode D4 is deactivated because the anode of this diode receives a voltage near to zero volt, then cannot conduct the current. The voltage across the inductors L1 and L2 can be shown by: Since the same inductor values are used, therefore, in all equations inductors L1 and L2 can be shown with L. in (1), 1, 2 presents the voltage across the inductors and shows the input voltage. Diode D7 is in forward-biased working state and the capacitor Co is charged. For the time interval that the power switch is deactivated, the diode D1 works in ON-mode and charges the capacitor C1.
Also, both inductors begin to discharge since a serial connection will be inevitable in this operational mode and the voltage across per inductor will be negative since a high value of the voltage drops across the capacitor C1 to charge it. In this working mode, both diodes D2 and D3 are deactivated. Also, diode D6 since receives a negative voltage at the cathode in comparison with the anode voltage that is connected to the positive pole of the capacitor C3 is activated and for the same reason, the diode D5 is deactivated. Diode D7 is in Off-mode since the cathode of this diode is connected to output voltage that always is greater than the input voltage. For this mode, since a serial connection forms between the inductors, the KVL through the input voltage and capacitor C1 can be presented by: Therefore, the voltage across the inductors can be calculated by (3): Since the total value of the voltage drops across an inductor in a switching circuit is zero based on the periodic operation, one can write: , Therefore, the voltage across the capacitor C1 can be written: This equation shows that the proposed converter can increase the input voltage (1 + ) times more than a conventional boost converter by adding a SL-cell. Furthermore, this configuration needs to currents with smaller ripple magnitudes in comparison with the current of the input inductor in classical boost converter. This subject will be analyzed in detail in the next subsection. ℎ voltage across the capacitors C2 and C3, one can write: And: VOLUME XX, 2017 Then, the voltages for the capacitors C2 and C3 can be obtained: Therefore, the gain of the converter can be obtained: This equation shows that the proposed converter can enhance the input voltage to 7.5 times for the duty ratio = 0.5 and closed to 20 times for the = 0.8. The greater voltages can be obtained by more cascaded SC-cells. A simple comparison easily can show that the gain of the proposed converter by considering the number of the used components numbers is acceptable and comparable with many of the recently designed boost converters in literature.

C: Current ripples for the inductors:
One of the most important features of the converter is achieving to limited ripples for the input current. It is essential for the long-life and reliability of the connected RES source especially if it is a PV module. Therefore, the calculation for this parameter should be presented. When the power switch is activated, the current of the inductors can be presented by (10, and 11): In (11), (∆ ) is the ripple of the inductors when the switch is in On-sate. Also, for the time intervals that switch is deactivated, the current ripple of the inductors can be found by (12, and 13): (∆ ) presents the ripple of the current for the inductors when the switch is deactivated. Also is a time period of the switching signal. Since the voltage across the capacitor C1 is approximately equal to the output voltage of the converter without considering the SC-cell, therefore one can change For the ideal working conditions, the average power absorbed by the load is the same with the generated power by the source. If a load like is considered as the load, the output power , equal to the input power can be calculated simply by (15): This equation can be written by considering that the current of the inductors is the same with current of the input source for the time interval that the switch is in ON-state. Therefore: The average amount of the inductor current can be calculated by (17): The maximum and minimum values of the inductor current can be found by (18) and (19) by considering the (11) and (17): These recent equations show that the minimum and maximum amplitude of the ripples for this converter is at least equal to the half of the ripples amplitude for the conventional boost converter. For Continuous Conduction operational Mode (CCM), , needs to be greater than zero. Therefore, for obtaining the boundary between CCM and Discontinuous Conduction Mode (DCM), inductor current can be determined from (20): In this equation is the switching frequency. The minimum value of the inductors and switching frequency of the proposed boost converter for operating in CCM is therefore: This equation easily can show that the selected inductor values are dependent to the duty ratio, the load value and the switching frequency.

D: Voltage ripple for output capacitor:
A fixed voltage with minimum value of the ripple should be presented at the output sides of the converter. This will prepare an appropriate working condition for the load block.
The output capacitor value and the voltage ripple across this capacitor will be investigated in this sub-section. It is possible to find the voltage variation across a capacitor by considering the currents of the capacitor. The capacitor electrical charge , can be calculated as (22) for the time interval that the 2 is charged: This equation has been written based on the fact that the current of the output capacitor for the ON-state time interval of the switch is equal to the . Therefore, the voltage ripple for this capacitor can be found as presented in (23): (24) shows that it is possible to calculate the value of the capacitor in terms of the output voltage ripple: The state of the current and voltage for the components of the boost converter is shown in figure 5.

E: Controller design:
By considering a load like , at the output points of the converter, for the time intervals that the power switch is activated, , the equation (26) can be written: With a simple placement in (26), one can write: In this equations, is the output resistance of the boost converter. Also, , and are the input current and output signal of a classic PI controller. Equation (27) indicates that the control signal of the controller has a direct relation with the value of the capacitor CO and the values of currents of the input source, inductors and the capacitor C2. By a simpler method, duty ratio that will be implemented to the power switch of the converter can be presented by (28): Figure 6 presents the controller scheme of the proposed converter and is obtained according to (28). The transform function of a PI controller ( ), can be written by (29): The closed-loop − function of the controller then can be written by: and are the proportional and integral coefficients of the PI controller and can be obtained as: In recent equation, and 0 are the damping factor and frequency.

F: DCM analysis:
To analysis the DCM working mode, all of the semiconductor devices should be considered in OFF-state. The reason is that, in DCM working mode, the currents of the inductors reach to zero and a voltage equal to zero can be measured across them. This state can occur after the second state in CCM. In this working mode that is not desired normally, the voltages for the capacitors is considered to be fixed. The currents of the inductors are important because for a boost converter, the load current is supplied by the inductors.
Receiving to the zero current in the inductors ( 1, = 2, = 0) means the inability to supplying the load and delivering the current and power to the load. Figure  7a presents the voltage and current waveforms for the inductors within a time period in DCM and figure 7b shows the state of the proposed converter for this operational mode. Since the currents for the inductors reach to zero, the diodes 1 and 4 go to off-state and no current can be established from the DC input source since the switch is in off-state. The current equations for the power diodes and the switch can be presented by the below equation: The current of the inductor in this mode is shown by 1, . The currents of the diodes 2 and 3 are zero since these diodes are activated in CCM and when the switch is activated. Therefore, one can write:

VL1
The current for the diode 4 is zero since this device is passing the current of the inductors: By disconnecting the loop current from the input side and by having a zero current for the output diode 7 , the currents for the capacitors in switched-capacitor block reach to zero. Therefor: As mentioned, this states occurs after the second working station in CCM and in this mode the switch is deactivated. The reflection of the diodes and the switch easily can show that the connection with the input source and the load will be disconnected and only the output capacitor exists to supply the load.

III. CALCULATION FOR COMPONENTS AND LOSSES, COMPARISON, AND DISCUSION
The components values in the circuit should be designed based on the working power value, and desired output voltage. Since the gain of the converter shows that can present a voltage closed to the 360 VDC at the output nodes by a 48 VDC input source, and in order to set up the circuit in the lab, a prototype with about 200 W, as the output power is analyzed. A simple calculation shows that a resistive load closed to the 650 Ω should be considered and a current near to the 550 mA is expected. The components values are presenting through the equations below.

A: Inductors values and input current calculations:
As mentioned, both of the inductors have the same value. For an acceptable current ripple for an inductor, a maximum variation of current ripples closed to 40 percent is considered. Duty cycle is considered to be = 0.5, and switching frequency is equal to f=50 kHz. Since the ripple is the same for the both inductors, through the (9), and (11): The inductor values can be shown by (43). (43) shows that for a larger switching frequency a smaller inductor can be used. On the other hand, for a limited ripple for the current, a larger inductor can be selected. Since this study plans to present the experimental test results, therefore 50 kHz as a reliable frequency is chosen and the obtained inductor value is proper.
The configuration of the proposed converter shows that the input current is the same with the total currents of the inductors 1 , 2 when the switch is activated and is equal with the currents of these inductors for time intervals that the switch is deactivated. (17) presents the average current of the inductors. For times the switch is in ON-state, by using the (11), the inductors are magnetized and the input current is obtained as: The input current for the time duration that the switch is deactivated by using the (13) A comparison between the input current stresses for the proposed converter and some of the presented similar converters by the researches in recent years is presented in Table II.

B: Capacitors values:
Equation (23) presents the relation between the output capacitor value and the voltage ripple. For a good converter, by a fixed output voltage, normally a ripple closed to 1 percent is considered. Therefore, the output capacitor can be selected as: This value shows that for a larger capacitor the output voltage will be more fixed. In practice, since many of the parasitic components are included in the account that normally in the theoretical investigations are ignored, a larger ripple value is expected. For that, instead of the calculated capacitor value in (49) As the same facts mentioned for other devices, a larger capacitor should be considered. Larger capacitor will fixed the voltage for the SC-cell, therefore 1 = 10 µ is selected. Table 1 presents all values for the elements:

B: Power losses calculations:
The efficiency of the converter is obtained by dividing the output power over the input power. The power losses in a converter structure can be divided into dynamic and switching losses. Dynamic losses can be calculated for all semiconductor devices and passive elements since they can transfer the current in the circuit while the switching losses occur only for power semiconductor devices including the switch and diodes. To find the dynamic losses, the rootmean-square (RMS) value of the currents for each element should be calculated.

• Power losses calculations for the switch:
The relation between the input and output currents can be presented by through equation (40): The current of the switch is the same with the input source current: The RMS current for the switch can be presented by: The real and RMS current for the diode 1 are : Therefore, the dynamic power loss can be calculated: The switching losses 1, ℎ for this diode can be calculated as below: 4, The power losses can be calculated for these diodes as below: (87)

• Power losses calculations for the inductors and capacitors:
To find the dynamic power losses for passive elements, the internal resistance of these components should be multiplied to the square of the RMS current of the components: The efficiency equation can be shown through:

D: Simulation results:
A group of the simulations have been done for the proposed converter. Figures 8 to 11 shows the results. In figure 8, the reaction of the inductors in SL-cell is investigated. The voltage across the inductors are reported in figure 8a and the currents of the inductors can be seen in figure 8b. These figures have been obtained for the elements in Table I. To verify the performance of the proposed converter, the circuit is tested under different working conditions. For example, the voltage and currents of the inductors are obtained with different values of the inductors. The values of the inductors are considered to be the same. In practice, it is hard to prepare similar inductors, and tolerance of about a maximum of 5 percent can be accepted. Figures 8c, 8d, 8e, 8f, 8g, and 8h present the voltage and current for the inductors with the same values, 5 percent differences, and 10 percent differences in values respectively. As can be seen, an ignorable difference for the current and almost the same voltage waveforms can be obtained and it does not degrade the performance of the circuit much.
Since the same values of the inductors are used, and both inductors are magnetized and demagnetized simultaneously, the voltage and current of the inductors are the same. For a 200 W prototype circuit, with 48 VDC and 360 VDC as the input and output voltages, a positive and negative voltage with 50 V magnitude and a ripple in current equal to 1 A are reported. A resistive load closed to 650 Ω is considered for the simulations and this means that a current equal to 0.55 A is expected at the output load. Figures 9a and 9b show the input and output voltages and currents respectively. Results show that all desired voltage and current values for the output side is obtained.  Figure 9b shows that an average current closed to 4.25 A is presented by the source. Also, the reaction of the converter under variable loads should be considered. For that, figure 9c is presented to show the performance of the converter in generating the different load currents for different load values. For example, for a load equal to 580 Ω, with about 360 VDC, a current closed to 615 mA is obtained. At the times = 0.1, and = 0.2 the second and third loads are involved at the output side. These loads are the same with the first load, therefore an enhancement for the output current is expected at these times. Figure 9c shows that the load current increases to 1.230 A and 1.84 A respectively at times = 0.1, and = 0.2.
In figure 10a, the currents of the power switch and the diode D1 are presented. An average current equal to 3.5 A and 0.75 A pass through the switch and diode D1 respectively. Total of both these currents follows from the input source and can be seen in figure 9b. The voltage across the drain-source pins of the switch is seen in figure 10b. This figure reports a maximum voltage stress closed to 140 V for this switch at the 360 VDC of the output voltage. Figure 11 presents the voltage across the diodes. This figure shows that when the switch is activated, diodes D2, D3, D5, and D7 are in Onstate and for the time intervals that the switch is deactivated, diodes D1, and D6 are in ON-state respectively. The correctness of the figure 5 can be proved through this figure.
The voltage states of the diodes D1, D2, and D3 are shown in figure 11a and the voltage waveforms for the diodes D5, D6 and D7 can be seen in figure 11b.

E: Comparison:
To evaluate the performance of the proposed converter, a group of the SC and SL-based boost converters are considered and the results of the comparison is presented in Table II. For the comparison, the number of the inductors, capacitors, diodes and switches and the voltage stress across the main switch and the gain of the proposed converters are examined. Based on the results, the converter in [31] has the most number of the inductors, and converters in [33], [36], and [37] include the most number of the capacitors. The proposed converter with the circuits in [33], [35], and [39] include only one power switch that is a proper feature for having a simpler controller circuit. However, the proposed converter and converter in [37] include the most number of the power diodes. Figure 12 illustrates the voltage gain curves for the converters in Table II. This figure shows that the gain of the proposed converter in all duty ratios, is more than converters in [30][31][32][33] and [35]. The converter in [34] has the most values of the voltage gain and closed to the gain of proposed converter in this study. Converters in [36] and [37] act as the voltage multiplier circuits and enhance the input voltage through a fixed switching frequency and duty ratio. Therefore, in figure 12, the gain of these converters are drawn fixed for all duty ratios. Circuit in [36] increases the input voltage by 6 times and converter in [37] enhances the source voltage by 4 times respectively. Also, the voltage stress curves for the converters in Table II are presented in figure 13. Where the converter in [35] presents a stress closed to the output voltage, other converters present smaller voltages and the best results can be obtained for the converters in [34], [33] and proposed converter respectively. The numbers in vertical axis should be multiplied to the output voltage at the same duty ratio to find the expected voltage magnitude.    Figure 14 shows the voltage stress across the output power diode. Results show that the maximum voltage stresses are reported for converters in [30][31][32], the middle stresses are belonging to the converters in [33] and [34] and the least stress is seen for the proposed converter.    Table II. Although in this table, the current stress equations are presented for the majority of the proposed topologies, since some of these equations are related to parameters such the switching times or current ripples in inductors that can be shown difficultly, the current of the topologies in [30] and [38] are shown. This figure shows the current ripple for duty ratios between 0.5 to 0.6 and as can be seen, the current stress is obtained between the ripples for converters in [30] and [38]. This shows that that the proposed converter can

IV. EXPERIMENTAL RESULTS
This section presents some of the results for the hardware tests. For that, a 200 W converter circuit was established in the laboratory according the elements value of the theoretical investigations and the simulation. A resistive load equal to 650 Ω is considered, and output voltage closed to 360 VDC is considered to be obtained for this load. The switching frequency is adjusted at the 50 kHz and the FQP34N20 type of the MOSFET is selected for the switch. This transistor has 35 mΩ between drain and source pins for the short-circuit working condition. Also the FRD type with IN5402 number is chosen for the all power diodes. The dynamic resistance for this type of the power diode is equal to 12 mΩ. The forward voltage for this diode based on the datasheet is equal to 1.2 V. Figure 16a presents the implemented prototype circuit for the proposed converter. Figure 16b shows the input and output currents of the converter. By the measured results, output current is a DC current with 550 mA magnitude. Input side current is divided between the switch and D1, and as mentioned in sections 2 and 3, around 4 A for the average current for the source side is expected. In test results, at the duty cycle neat to 50 percent at the Fs=50 kHz, the expected current is obtained. Figure 16c shows the state of the power switch and illustrates that a voltage near to 140 V drops across the drain-source pins with a current ripple from zero to 8A. An average current closed to 4A is obtained. The frequency and duty ratio of these waveforms as can be seen are the same with the desired values. The current waveforms and voltage drops across the inductors are similar and can be seen in figure 16d. A voltage with peak to peak 120 V is obtained and a current with maximum 3.2 A and minimum around 2 A is reported. These result has a proper overlap with the simulation results since approximately a same current is obtained and about 20 V more than simulation results for the inductors voltages are reported. This can be because of the internal resistance of the inductors that were not considered for the theoretical analysis and also for the simulation tests. Figure 16e presents the voltage across the diodes D2 and D3. These two diodes are in series with the inductors and activated and deactivated simultaneously. To show the reaction of the diodes in SC-cell, figure 16f is presented. According to the simulation results, these diodes are in ON and OFF-states asynchronously and for any time interval only one of these diodes is activated. Results show that a voltage with maximum 50 V peak to peak is obtained. By considering that the diodes and capacitors in SC-cell are the same, the voltages across these devices also should be the same. Figure 16f shows that the diodes work alternatively and voltages across the anode-cathode pins is the same. Figure 16g Figure 16j illustrates the capability of the converter on the output side when the loads are changing. Two similar 580 Ω resistive loads through a breaker are connected at the output side and the ability of the converter for presenting the necessary currents for these loads is investigated. This figure shows that the output current approximately is doubled when the second load is applied. Figure 17 presents the efficiency diagram for the converters in Table II. For that, the efficiency of the converters for power values from zero to 200 W is read. Also, the efficiency curve for the proposed converter is added. Figure 17. Efficiency of the converters in Table 2 and proposed converter.
According to the results of this figure, the converters in [30], [34], and [35] present the maximum efficiencies for powers less than 125 W. After this power value, the proposed converter with converters in [30] and [34] present the highest efficiencies. Converters in [35] and [38] are suitable for lowpower applications and the efficiency of the converter in [36] for higher power values is more and make the converter proper for high-power applications. For drawing the figure, authors tried to present curves with the closest input voltages and working conditions for all of the converter for a fair comparison. The efficiency of the proposed converters is more than 90 percent for all output power values and can present higher efficiencies for higher power values. For powers below 150 W, this converter can present higher efficiencies in comparison with a large part of these converters and for powers more than 150 W, it can present the highest efficiency. This property is important since can prepare the converter for high-power applications. Figure 18 presents the efficiency diagram for the proposed converter with different input voltages and 360 VDC as the fixed output voltage for output power of 200 W. As was predictable, with an enhancement in input voltage, the converter acts more efficient and this is because of fewer differences between the input and output voltages. For fewer input voltages longer duty cycles are needed to be applied to the switch and diodes, and since the semiconductors have most of the losses, the efficiency is less.

IV. CONCLUSION
A boost converter with a switched-inductor and a switchedcapacitor topology is presented in this study. By adding an inductor to the conventional boost converter and replacing the locations of the inductors, transistor and diode, a new configuration for the proposed boost converter is presented. The most important features of this configuration is enhancing the input voltage with more gain ((1 + ) times greater) and decreasing the input current ripple in comparison with the classic boost converters. All these specifications will be gained only by an extra inductor and three power diodes for the SL configuration. Since diodes are used instead of the transistors in SL-side, there is no need for the control circuit. For the next step, the proposed SC-cell is introduced with a simple configuration including two capacitors and two diodes. This block enhances the voltage at the output of the converter since for the duty ratio closed to 0.5, a voltage equal to 7.5 times greater than the input voltage is obtained. Also, with the proposed SC-cell for the duty ratio closed to 0.8, approximately 19 times greater than the input voltage is obtained. Since working under the greater duty ratios can lose more powers because of the longer time to pass current for obtaining more gains, by considering that the SC-cell has only four devices, a cascaded configuration can be considered for receiving to high efficiencies. This configuration is suitable for grid integration Photovoltaic (PV) applications, since the generated voltages of the PV arrays is limited and needs to enhance more. A large part of the voltages and currents of the converter components are reported theoretically, in simulation and practically. A laboratory-scaled 200 W prototype is presented and confirms the theoretical results.