Dual-Band Circularly Polarized Omni-Directional Biconical Antenna With Double-Circular Parallelepiped Elements for WLAN Applications

This research proposes a novel dual-band (2.45/5.80 GHz) omnidirectional circularly polarized (CP) biconical antenna with double-circular parasitic parallelepiped elements for wireless local area network (WLAN) applications. The proposed dual-band CP antenna scheme consisted of a biconical radiating structure surrounded by inner- and outer-circular parallelepiped elements that convert linearly polarized electric fields into CP fields. Simulations were performed to optimize the antenna parameters, and an antenna prototype was fabricated and experiments were conducted. The measured impedance bandwidths (IBWs) were 44.4% (1.84 – 2.89 GHz) and 4.56% (5.73 – 5.99 GHz) for the lower- (2.4 GHz) and upper-frequency (5.80 GHz) bands, respectively. The corresponding 3-dB axial ratio bandwidths (ARBWs) were 11.22% (2.27 – 2.54 GHz) and 10.49% (5.6 – 6.2 GHz). The radiation patterns of the dual-band antenna scheme were omnidirectional left-hand circular polarization, with the measured antenna gains of 3.2 dBic and 8.5 dBic at 2.45 and 5.80 GHz, respectively. The simulated and measured results were reasonably agreeable. Despite the narrow IBW and ARBW for the upper-frequency band, the bandwidths adequately covered the target upper frequency band, rendering the proposed CP omnidirectional biconical antenna scheme operationally suitable for WLAN applications. Furthermore, the novelty of this research lies in the use of a biconical radiating structure augmented with double-circular parasitic parallelepiped elements to realize circular polarization for dual-band WLAN applications.


I. INTRODUCTION
The advancement in modern wireless communication The advantages of circularly polarized (CP) antennas include versatility, lower multipath loss, and higher The associate editor coordinating the review of this manuscript and approving it for publication was Giorgio Montisci . penetration, unlike the linearly polarized (LP) antennas which suffer from multipath loss and low penetration [1]. As a result, CP antennas are used in a variety of applications, including global navigation satellite systems, radar, radio frequency identification (RFID), and sensor systems [2]- [5]. In addition, CP antennas with multi-band coverage require fewer antenna components [6]- [9].
Furthermore, recent years have witnessed renewed interest in biconical antennas due to their straightforward design, ease of fabrication, and multiband coverage capability [10], [11]. In [12], wideband biconical antennas were proposed and experimented. A compact dual-band (1.575/2.45 GHz) LP/CP antenna scheme for outdoor/indoor applications in [13] was proposed, but it suffered from narrow impedance bandwidth (IBW). In [14], a dual-band (3.0/5.5 GHz) monopole antenna with frequency-selective surface-based corner reflector effectively achieved both frequency bands but failed to generate circular polarization. A dual-band (2.45/5.2 GHz) monopole antenna with multiple modified substrates for WLAN applications was proposed in [15]. Due to the standing wave current distribution along the longitudinal direction, the radiation pattern of conventional biconical antennas is thus of linear polarization [16].
As a result, a low-profile wideband CP circular patch antenna with two monopolar modes connected to a modified ground plane by conductive vias [17] was proposed to generate omnidirectional circular polarization. In [18], a patch antenna with vortex slots and shorting vias could achieve circular polarization. However, these CP antennas [17], [18] partially covered dual-band operating frequencies.
With the growing demand for dual-band CP antennas, a dual-band (2.45/5.80 GHz) antenna with adjustable stubs was proposed, and the antenna could achieve circular polarization but suffered from narrow axial ratio bandwidth (ARBW) in the lower-frequency band [19]. In [20], a dual-band (3.1/4.7 GHz) CP split-ring resonators with square slot was proposed, but it suffered from narrow ARBW. A dual-band CP planar monopole antenna in [21] could cover WLAN/Wi-Fi/Bluetooth/WiMAX frequency spectra but failed to achieve an omnidirectional radiation pattern.
Theoretically, an LP wave can be converted into a CP wave by using a wave polarizer [22], [23]. As a result, an omnidirectional CP antenna could be realized by incorporating a wave polarizer into an omnidirectional LP antenna. Specifically, this research proposes a novel dualband (2.45/5.80 GHz) CP omnidirectional antenna scheme for WLAN applications by integrating a biconical radiating structure with a wave polarizer consisting of double-circular (inner and outer circles) parasitic parallelepiped elements. The biconical structure at the center is surrounded by double-circular parasitic parallelepiped elements which are uniformly and angularly arranged along the circumference of the biconical structure to convert an LP wave into a CP wave.
In this study, simulations were performed using CST Studio Suite and a prototype antenna was fabricated. Experiments were performed and results compared with the simulations. The performance metrics of the dual-band CP omnidirectional biconical antenna with double-circular parasitic parallelepiped elements included the impedance bandwidth (|S 11 | ≤ −10 dB), axial ratio (AR ≤ 3 dB), radiation pattern, and antenna gain. The research novelty is the application of a biconical radiating structure and doublecircular parasitic parallelepiped elements to realizing circular polarization for dual-band WLAN applications. This paper is organized as follows. Detailed antenna configurations are discussed in Section II. The parametric study is carried out in Section III. Simulated and measured results are presented in Section IV. Finally, conclusions are shown in Section V. Figure 1(a) shows the proposed biconical antenna with the half-angle of the bicone of θ hc , where θ hc is calculated from the impedance (Z c ) of an infinite biconical antenna (equation (1)).

II. ANTENNA CONFIGURATION
where η is the impedance of free-space (=376.73 ) and θ hc is the half-angle of the bicone. A half-angle of approximately 67 • yields a 50 match. Since infinitely long biconical elements are impossible to realize, the length of the bicone must be truncated. Given that the diameter of the biconical antenna is approximately one-wavelength (1.0λ) for the target lower-frequency band (2.40 GHz) [24], the height of the biconical antenna (H ) is thus λ/2 of the lower-frequency band. In this research, the optimal parametric dimensions of the biconical antenna are as follows: θ hc = 71 • , H = 40 mm, and R 3 = 56 mm. Figure 1(b) shows the geometry of outer and inner parasitic parallelepiped elements functioning as the wave polarizer [22], [23] of the proposed CP biconical antenna scheme. The parallelepiped elements are of aluminum material.
Assuming an LP wave traveling in the x-direction with a 45 • polarization relative to the positive y-axis, the electric field of the LP wave can be resolved into E y and E z components. Since the parasitic parallelepiped elements and air gap behave like a wave polarizer, the two electric field components (E y and E z ) would travel at different speeds as the LP wave passes through the polarizer, resulting in a 90 • phase difference.
By varying the outer and inner parallelepiped element thicknesses (d 1 and d 2 ), a 90 • phase difference could be achieved, giving rise to circular polarization given that |E y | = |E z |. As a result, an omnidirectional CP antenna could be realized by angularly encircling an omnidirectional LP radiating source with a polarizer (i.e., double-circular parasitic parallelepiped elements).
In this research, the radiating source is a biconical structure of copper material. Since the electric field generated by the biconical structure is of vertical linear polarization, tilted parasitic parallelepiped elements were utilized to induce the polarizer effect (i.e., circular polarization). Figure 1(b) illustrates the perspective view of the proposed dual-band CP omnidirectional biconical antenna scheme. Figure 1(c) shows the arrangement and displacement (from the center of the biconical structure) of the inner and outer parallelepiped elements around the biconical radiating structure. The parasitic parallelepiped elements are uniformly arranged along the circumference of the biconical structure with an angular interval of 360 • /N , where N is the number of inner and outer parallelepiped elements. Figure 1(d) shows the front view of the dual-band CP biconical antenna scheme with double-circular parallelepiped elements, with the feeding point at the contact point between the bicone.

III. PARAMETRIC STUDIES
To optimize the antenna parameters, simulations were performed using CST Studio Suite. In this study, the target lower-and upper-frequency bands of the dual-band CP antenna scheme were 2.40 -2.485 and 5.725 -5.875 GHz. With N = 4 (i.e., the inner-and outer-circle parallelepiped elements were four each), the IBW (|S 11 | ≤ −10 dB) was 1.99 -5.83 GHz, covering both lower-(2.45 GHz) and upper-frequency bands (5.80 GHz). The wide IBW could be attributed to limited number of inner-and outercircle parasitic parallelepiped elements (i.e., N = 4 each). However, the AR was above 3dB (AR > 3 dB) for both lower-and upper-frequency bands.
With N = 8, the IBW was 1.95 -3.04 GHz and 3.46 -5.72 GHz for the lower-and upper-frequency bands, respectively. An impedance mismatch (|S 11 | > −10dB) occurred in the upper-frequency band. The ARBW (AR ≤ 3 dB) was 3.90% at the lower-frequency band, covering 2.51 -2.61 GHz. However, the AR was substantially larger than 3 dB (AR > 3 dB) for the upper-frequency band.
With N = 16, the IBWs and ARBWs of the lowerfrequency band were 1.84 -2.88 GHz and 2.31 -2.57 GHz respectively, while those of the upper-frequency band were 5.72 -6.01 GHz and 5.59 -6.05 GHz respectively, achieving circular polarization for both bands. With N = 4, the proposed biconical antenna scheme failed to achieve omnidirectional radiation pattern for both lower-and upper-frequency bands. With N = 8, the proposed antenna scheme achieved omnidirectional radiation pattern in the lower-frequency band, but the radiation pattern was non-omnidirectional in the upper-frequency band. With N = 16, the radiation patterns of the lower-and upper-frequency bands were omnidirectional. As a result, the optimal numbers of inner-and outer-circle parasitic parallelepiped elements (N ) were 16 each. Figure 4(a) shows the simulated |S 11 | and AR of the proposed dual-band CP biconical antenna scheme under different outer-circle parallelepiped widths (w 1 ): 4, 6, and 8 mm. With w 1 = 4 mm, the simulated IBWs (|S 11 | ≤ −10 dB) were 1.84 -2.9 GHz and 5.6 -5.94 GHz for the lower-and upper-frequency bands, respectively. The corresponding simulated ARBWs (AR ≤ 3 dB) were 2.36 -2.54 GHz and 5.71 -5.92 GHz.
With w 1 = 6 mm, the impedance matching remained relatively unchanged for the lower-frequency band, while that of the upper-frequency band deteriorated (from −30 dB to -18 dB). The 3-dB AR remained relatively unchanged for the lower-frequency band, while that of the upper-frequency band improved, resulting in wider ARBW (5.59 -6.05 GHz).
With w 1 = 8 mm, the impedance matching of lowerfrequency band significantly improved (|S 11 | < −30dB), while that of the upper-frequency band deteriorated and failed to cover the target upper-frequency band. The 3-dB AR remained relatively unchanged for the lower-frequency band, while that of the upper-frequency band deteriorated (AR > 3 dB), with ARBW partially covering the upper-frequency band (5.56 -5.72 GHz). As a result, the optimal w 1 was 6 mm. Figure 4(a) shows the simulated |S 11 | and AR of the proposed dual-band CP biconical antenna scheme under different outer-circle parallelepiped widths (w 1 ): 4, 6, and 8 mm. In comparison with w 1 = 4 mm, given w 1 = 6 mm, the impedance matching remained relatively unchanged for the lower-frequency band, while that of the upper-frequency band deteriorated. The AR remained relatively unchanged for the lower-frequency band, while that of the upper-frequency band improved, resulting in wider ARBW.
With w 1 = 8 mm, the impedance matching of lowerfrequency band significantly improved, while that of the upper-frequency band deteriorated and failed to cover the target upper-frequency band. The AR remained relatively unchanged for the lower-frequency band, while that of the upper-frequency band deteriorated, with ARBW partially covering the upper-frequency band. As a result, the optimal w 1 was 6 mm. Figure 4(b) shows the simulated |S 11 | and AR of the proposed antenna scheme under variable inner-circle parallelepiped widths (w 2 ): 0.5, 1, and 1.5 mm. The simulated results were almost identical to those of w 1 (Figure 4(a)), and the optimal w 2 was 1 mm. Figure 4(c) shows the simulated |S 11 | and AR under different outer-circle parallelepiped thickness (d 1 ): 17, 19, and 21 mm. With d 1 = 17 mm, the AR of the upperfrequency band was excessively high, failing to achieve circular polarization. With d 1 = 19 mm, the impedance matching slightly improved for the lower-frequency band (|S 11 | = −28 dB), while that of the upper-frequency band deteriorated. The AR were below 3 dB for both lower-and upper-frequency bands. With d 1 = 21 mm, the impedance matching of the lower-frequency band significantly improved (|S 11 | < −43 dB), while that of the upper-frequency band deteriorated and shifted to higher frequency, failing to cover the target upper-frequency band. The AR remained relatively unchanged for the lower-frequency band, while that of the upper-frequency band noticeably deteriorated, with ARBW failing to cover the target upper-frequency band. The optimal d 1 was thus 19 mm. Figure 4(d) illustrates the simulated |S 11 | and AR under variable inner-circle parallelepiped thickness (d 2 ): 9, 10, and 11 mm. The results showed that varying d 2 had minimal impact on the impedance matching of the proposed antenna scheme. Meanwhile, varying d 2 minimally affected the AR of the lower-frequency band, vis-à-vis that of the upper-frequency band. The optimal d 2 was 9 mm since the ARBW covered the target dual-band frequencies.
Figures 5(a)-(b) show the simulated |S 11 | and AR under variable outer-circle parallelepiped element lengths (l 1 ): 0, 15, 25, 42, and 62 mm. In Figure 5(a), with l 1 = 0 mm, the impedance matching was achieved at 2.7 GHz and 4.4 GHz, which were outside the target dual-band frequencies (2.45/5.80 GHz). The impedance matching improved and moved closer to the target frequencies as l 1 increased from 15 mm to 42 mm. An impedance mismatch occurred when l 1 = 62 mm.
In Figure 5(b), with l 1 = 0, 15, and 25 mm, AR was above 3dB for both lower-and upper-frequency bands. With l 1 = 62 mm, AR was higher than 3 dB in the upper-frequency band. Thus, the optimal l 1 was 42 mm as AR was below 3 dB for both lower-and upper-frequency bands.
Figures 5(c)-(d) illustrate the simulated |S 11 | and AR under variable inner-circle parallelepiped element lengths (l 2 ): 0, 5, 10, and 13.3 mm. In Figure 5(c), except for l 2 = 10 mm in the lower-frequency band, varying l 2 had minimal impact on the impedance matching of the proposed antenna scheme. In Figure 5(d), varying l 2 had minimal impact on the AR in the lower-frequency band, while the AR of the upperfrequency band improved as l 2 increased. The optimal l 2 was 13.3 mm as the IBW and ARBW covered both lower-and upper-frequency bands. Meanwhile, given H = 38.5 mm, the ARBW covered 2.37 -2.57 GHz, while that for the upper-frequency band of 5.12 -5.3 GHz was far below the target frequency band. With H = 40.5 mm, the AR was below 3 dB for both the target lower-and upper-frequency bands. However, with H = 42.5 mm, the AR of the upper-frequency band was well above 3 dB, failing to achieving circular polarization. The optimal H was thus 40.5 mm. Figure 6(b) shows the simulated |S 11 | and AR under different displacements of outer-circle parallelepiped elements (R 1 ): 88, 90, and 92 mm, where R 1 is the radius between the outermost edge and the center. With R 1 = 88 mm, the IBW for the lower-and upper-frequency bands covered 1.87 -2.83 GHz and 5.94 -6.22 GHz. With R 1 = 90 mm, the IBW covered both the target lower-and upper-frequency bands. However, the IBW for the upper-frequency band covered 5.05 -5.81 GHz for R 1 = 92 mm, partially covering the target upper-frequency band.
Meanwhile, with R 1 = 88 mm, the ARBW of 2.29 -2.55 GHz covered the target lower-frequency band, while that of the upper-frequency band of 5.51 -5.62 GHz was below the target frequency band. With R 1 = 90 mm, the AR was below 3 dB for both the target lower-and upperfrequency bands. With R 1 = 92 mm, the AR of the upperfrequency band was well above 3 dB, failing to achieve circular polarization. As a result, the optimal R 1 was 90 mm. Figure 6(c) shows the simulated |S 11 | and AR under different displacements of inner-circle parallelepiped elements (R 4 ): 30.5, 31.5, and 32.5 mm. R 4 is the radius from the center to the outer edge of the inner-circle parallelepiped elements. The results showed that R 4 had minimal effect on the impedance matching of the proposed dual-band CP antenna scheme. The AR for the lower-frequency band was below 3 dB, independent of R 4 . For the upper-frequency band, with R 4 = 30.5 mm, the AR was above 3 dB. For R 4 = 31.5 mm, the ARBW covered the target upperfrequency band. With R 4 = 32.5 mm, the ARBW were outside of the target frequency band. The optimal R 4 was thus 31.5 mm. Figure 6(d) depicts the simulated |S 11 | and AR under variable radii of the biconical structure (R 3 ): 53, 56, and 59 mm. The impedance matching improved with increase in R 3 , but |S 11 | shifted to lower frequency as R 3 increased.  For the upper-frequency band, the IBW, given R 3 = 53 and 59 mm, failed to cover the target frequency band. With R 3 = 56 mm, the IBW covered both target frequency bands. The AR for the upper-frequency band, given R 3 = 53 and 59 mm, were well above 3 dB. Meanwhile, with R 3 = 56 mm, the ARBW covered the target upper-frequency band. The optimal R 3 was 56 mm. Figure 6(e) shows the simulated |S 11 | and AR under different angles of the outer-and inner-circle parallelepiped elements (α 1 , α 2 ): 40 • , 47 • , and 54 • . The impedance matching for the lower-frequency band generally improved with increase in α 1 and α 2 , but that for the upper-frequency band deteriorated as α 1 and α 2 increased. The AR for the lower-frequency band was below 3 dB, independent of α 1 and α 2 .
The parametric studies indicate that the position, size and tilt angle of the parasitic parallelepiped elements play a crucial role in the LP-to-CP conversion and the antenna performance, including the impedance and AR bandwidths, omnidirectionality, and gain. Specifically, the size of the parasitic parallelepiped elements (i.e., d 1 , d 2 , l 1 , and l 2 ) affect the AR bandwidth. Meanwhile, the position (R 1 and R 4 ) and the tilt angles (α 1 and α 2 ) of the parasitic parallelepiped elements affect the IBW and AR bandwidths and omnidirectionality. Besides, non-uniform distances between the parasitic elements also result in non-omnidirectional radition pattern in the xy plane. By comparison, at 5.8 GHz, the impedance and AR bandwidths are more sensitive to the arrangement of inner parasitic parallelepiped elements, vis-à-vis at 2.4 GHz

IV. SIMULATED AND MEASURED RESULTS
To verify, a dual-band CP omnidirectional biconical antenna prototype was fabricated and experiments conducted. Figure 7(a) depicts the prototype of the proposed dualband CP omnidirectional biconical antenna with doublecircular parasitic parallelepiped elements. Figure 7(b) shows the top view of the antenna prototype where the outer and inner parasitic elements were arranged around the biconical antenna. In the antenna prototype, closed-cell extruded polystyrene foam (i.e., Styrofoam) was used as the divider between the parasitic elements with a tilt angle of 47 • . The dielectric constant of Styrofoam (ε r = 1.1) is close to that of air (ε r = 1.0006) [25]. Fig. 7(c) shows the biconical antenna structure, consisting of welded tapered copper sheets of 1 mm in thickness each. The biconical antenna is fed by a coaxial cable at the center of the structure. Fig. 7(d) shows the outer and inner parasitic elements (without the biconical structure) with Styrofoam between the parasitic elements. The |S 11 |, AR, radiation pattern, and antenna gain were measured using an HP8520C network analyzer in an anechoic chamber. The ETS-Lindgren Model 3100 Series Conical Log Spiral antennas were used to verify left-hand circular polarized (LHCP) and right-hand circular polarized (RHCP) radiations.    Figure 9 compares the simulated and measured ARs of the dual-band CP omnidirectional biconical antenna scheme at θ = 90 • . The simulated ARBWs (AR ≤ 3dB) were 2.31 -2.57 GHz) and 5.60 -6.05 GHz for the lower-and upper-frequency bands respectively, while the corresponding measured ARBWs were 2.27 -2.54 GHz and 5.60 -6.22 GHz. The simulated and measured results were reasonably agreeable, with discrepancies in the upper-frequency band attributable to non-uniformity of the parallelepiped element angles (α 1 , α 2 ) and displacements (R 2 and R 5 ) along the biconical circumference during the antenna fabrication. In this research, there were 16 parasitic parallelepiped elements each for inner and outer circles. Figure 10 illustrates the simulated and measured antenna maximum gains at the lower-frequency and upper-frequency bands of the dual-band CP omnidirectional biconical antenna scheme. The simulated and measured gains were agreeable. The measured antenna gains were higher than 2.3 dBic for both the lower-and upper-frequency bands, with the maximum gains of 3.2 dBic and 8.5 dBic at the center frequencies of the lower-(2.45 GHz) and upper-frequency (5.80 GHz) bands. The slightly lower measured gains were attributable to the antenna fabrication. Figures 11(a)-(d) illustrate the simulated electric field of the proposed CP omnidirectional biconical antenna scheme (front view) at 2.45 GHz at t = 0, T /4, T /2, and 3T /4, respectively. At t = 0, the electric field flowed upward, and at t = T /4, the electric field shifted 90 • and traveled rightward. At t = T /2, the electric field flowed downward, with a 90 • phase shift vis-à-vis at t = T /4. At t = 3T /4, the direction of the electric field was leftward. The electric field of the proposed CP omnidirectional biconical antenna scheme rotated counterclockwise (i.e., LHCP).   At 5.80 GHz, the HPBWs of the simulated and measured LHCP radiation patterns in the yz-plane were 80 • and 80 • respectively, as shown in Figure 13(b) on the lefthand side. The simulated and measured radiation patterns were reasonably agreeable. the XPL was around 20 dB. In Figure 13(b) on the right-hand side, the simulated and  measured radiation patterns in the xy-plane were of LHCP omnidirectionality, and the XPL was around 20 dB. Table 2 compares the parameters and performance of the existing antenna schemes and the proposed dual-band CP omnidirectional biconical antenna with double-circular parasitic parallelepiped elements. Specifically, despite more compact than the proposed antenna scheme, the planar, lowprofile, dual-band and dual-polarized antenna in [13] yields unidirectional pattern with single-band circular polarization. Besides, the impedance and AR bandwidths are narrower than those of the proposed antenna scheme. In [15], although smaller in size, the substrate-integrated suspended line highefficiency monopole antenna radiates bidirectional pattern in dual-frequency band.
In [17] and [18], the CP omnidirectional antennas achieve wider impedance and AR bandwidths, but the antennas can radiate in single-frequency band with antenna lower gain than the proposed antenna scheme. Despite smaller in size and wider impedance bandwidth, the antenna in [19] has bidirectional radiation.
In [20], although smaller, the dual-band CP split ring resonators loaded square slot antenna fails to achieve omnidirectional radiation pattern, with narrow AR bandwidth. In [21], the dual-band CP planar monopole antenna has not radiated omnidirectionally despite smaller in size.
By comparison, the proposed antenna scheme can achieve circularly polarized omnidirectionality in dual-frequency band (2.45/5.80 GHz) in spite of the larger electrical dimension.

V. CONCLUSION
This research proposed a novel dual-band 2.45/5.80 GHz CP omnidirectional biconical antenna with double-circular parasitic parallelepiped elements for WLAN applications. In the antenna development, the double-circular (inner and outer circles) parasitic parallelepiped elements were uniformly and angularly arranged surrounding the biconical radiating structure, and the double-circular parallelepiped elements functioned as a wave polarizer that converts LP omnidirectional electric fields from the biconical radiating structure into omnidirectional CP fields. The target lowerand upper-frequency bands of the proposed dual-band CP omnidirectional antenna scheme were 2.4 -2.485 and 5.725 -5.875 GHz. In the study, simulations were performed to optimize the antenna parameters, and an antenna prototype was fabricated and experiments were conducted. The simulated IBWs were 44.0% (1.84 -2.88 GHz) and 4.62% (5.71 -5.98 GHz) for the lower-and upper-frequency bands, respectively. The corresponding measured IBWs were 44.4% (1.84 -2.89 GHz) and 4.56% (5.73 -5.99 GHz). The simulated ARBWs were 10.65% (2.31 -2.57 GHz) and 7.73% (5.6 -6.05 GHz), while the measured ARBWs were 11.22% (2.27 -2.54 GHz) and 10.49% (5.6 -6.2 GHz), respectively. The simulated and measured radiation patterns of the proposed CP biconical antenna scheme were LHCP omnidirectional, with the measured antenna gains of 3.2 dBic and 8.5 dBic at 2.45 and 5.80 GHz, respectively. Despite the narrow IBW (4.56%) and ARBW (10.49%) for the upperfrequency band, the bandwidths adequately covered the target upper frequency band (5.725 -5.875 GHz), rendering the proposed CP omnidirectional biconical antenna scheme operationally suitable for WLAN applications. Nonetheless, to address the narrow IBW, subsequent research would experiment with different geometries of the radiating structure (i.e., different structures other than the biconical). In addition, to tackle the relatively narrow ARBW, future research would experimentally vary the dimensions and angles of the inner-and outer-circle parasitic parallelepiped elements. The material of the parallelepiped elements could also be altered from aluminum to dielectric resonator materials to realize wider ARBW.