A Wearable Circularly Polarized Antenna Backed by AMC Reflector for WBAN Communications

A new flexible circularly polarized (CP) wearable antenna is proposed for Wireless Body-Area Network systems (WBAN) operating at 5.8 GHz. The circular polarization is enabled by combining a microstrip-line monopole feed and an inverted L-shaped conformal metal strip extended from a coplanar waveguide. The proposed antenna shows satisfactory performance in terms of gain and specific absorption rate (SAR) at a separation of 12 mm from a human body model. To minimize body-antenna separation, a flexible 2×2 artificial magnetic conductor (AMC) was used as a reflector to achieve good performances in terms of gain and SAR. The total footprint of the proposed antenna is only 34.4 × 34.4 mm2 (0.384 λ0 2) board of semi-flexible Rogers RT-Duroid 5880 substrate. The final design provides a peak gain of 7.6 dBi and an efficiency of 94.7% when worn on the body. Furthermore, evaluation results indicate that the maximum SAR level decreased by up to 20.42% in comparison with the CP antenna without the AMC. Full-wave EM simulation and experimental results are performed, both in free space and proximity to the human body under different bending scenarios. Overall, the proposed antenna performance has been shown to be robust to structural deformation along the x-axis in comparison to previously reported designs. These features demonstrate that the proposed antenna is a strong candidate for off-body wearable applications.


I. INTRODUCTION
Recently, the interest in wireless body area networks (WBANs) has been rising significantly in the development of flexible and wearable devices [1]. Additionally, antennas integrated into wearable devices have attracted strong interest due to their diverse applications in the personal entertainment, health care, and defense sectors [2]. Moreover, due to the growth in wireless sensors in biomedical applications, the demand for high-performance compact flexible wearable antennas has dramatically increased, particularly those capable of supporting efficient off-body communications [3]. Since wearable antennas operate within the vicinity of the body, the loading effect of lossy tissue and structural deformation such as bending and crumpling, must be considered in the antenna design process [4]. Furthermore, wearable antennas are subject to several shape distortions due to the uncertain form of the material surface during the wearer's daily activities. As a result, the scattering due to the body motion can modify the reflected polarization of the wave, causing polarization disparity [5]. Thus, wearable antennas with robust performance that solve problems in terms of influence of the lossy dielectric human tissue body, deformation, and polarization mismatch, are suggested for WBAN applications [6]. Several off-body communication antennas, with and without the integration of flexible/textile materials, have been proposed for wearable applications in the 2.4/5.8 GHz industrial, scientific, and medical (ISM) bands. Patch antennas [7], planar monopoles [8], and slot antennas [9], have been proposed for conventional wearable antennas applications. Hence, various antenna designs have been also introduced to achieve enhanced performance and provide multiple functionalities [10]- [16]. Specifically, slot antennas based on substrate integrated waveguide (SIW) technology, exhibiting either single or dual-band operation, have been discussed in [10]- [12]. Similarly, fractal [13]- [14], inverted-F [15], and textile antennas [16] have also been suggested to improve SAR levels when placed in close proximity to the human body. On the other hand, to minimize the human body effect, an isolating layer is used as a backing shield to the wearable antenna such as artificial magnetic conductor (AMCs) [17]- [18], electromagnetic bandgap (EBG) [19]- [20], high impedance surfaces (HISs) [21]- [22], and electric ring resonators (ERR) [23]. Therefore, radiation patterns toward the human body surface are reduced with good improvements in terms of gain and front-to-back ratio (FBR) are achieved [24]- [25]. It is noticed that most of the abovementioned antennas are linearly polarized (LP), which could result in a limitation for the function activities due to the polarization mismatch losses caused by the human body movements. The use of wearable circularly polarized (CP) antenna is the best way to mitigate this problem. For instance, a compact circularly polarized (CP) integrated filtering antenna is reported in [26] for wearable biotelemetric devices. In [27], a circularly polarized button antenna for 5 GHz WBAN applications has been studied. A compact and high-efficiency CP antenna for body-centric applications has been reported in [28]. However, the most reported CP antennas applied for WBAN applications are based on rigid substrates, which cannot exhibit conformal characteristics. In this area, few research works on CP wearable antennas have been reported [29]- [32]. The multilayer flexible CP antenna is introduced in [29]. In [30], a costly CP antenna has been suggested by using flexible polydimethylsiloxane (PDMS) as a substrate. In [31], a bulky circularly polarized wearable DRA has been also investigated for off-body communication. In addition, a wearable textile CP antenna was designed in [32] for location tracking purposes. Significant frequency shifts occur due to structural deformation resulted in an increase of beam width with a decrease in the gain. Moreover, these antennas have characteristics in common that limit performance, such as low efficiency, narrow bandwidth, frequency shifts, and poor front-to-back ratio (FBR) under different bending conditions, and their complex excitation mechanism made the design uncomfortable to the user. Therefore, a flexible CP antenna with significant performance and bending robustness capabilities are highly desirable for integrating into wearable devices. In this paper, a flexible CP wearable antenna backed by an AMC array structure is designed and investigated for WBAN applications. The proposed antenna consists of a coplanar waveguide (CPW)-fed microstrip line monopole antenna. An inverted L-shape parasitic strip from one side of the coplanar ground planes has been designed to achieve the CP capability at 5.8 GHz. Moreover, the monopole antenna is backed by a compact 2×2 AMC array, to reduce the backward electromagnetic radiation towards the human body. The proposed antenna system is thinner than the earlier reported ones. Besides, it provides good performances in terms of SAR and gain. The antenna has been fabricated and tested when placed in different places of the human body, with considering different bending scenarios. The remainder of this paper is structured as follows: Section II describes the antenna design. Section III deals with the implementation of the antenna with a novel array of periodic AMC model. Section IV analyzes the effects of human body loading on the performance of the integrated antenna under various scenarios, including antenna properties, and SAR levels. Summary and Conclusions are given in section V.

II. ANTENNA DESIGN AND ANALYSIS
In this section, the design process of the proposed CP antenna is investigated. In addition, numerical simulations and experimental data of S parameters, total efficiency, realized gain, and radiation patterns are examined.

A. ANTENNA DESIGN
In this study, the targeted application is specifically designed to operate at 5.8 GHz of the ISM band which can easily fit any wearable and conformal wireless device. The application is intended for health monitoring of medical status via wearable sensors, which is placed on a different human body part. The antenna topology consists of a coplanar waveguide CPW-fed CP monopole antenna with dimensions of L s ×W s . The antenna dimensions are determined in free space wavelength at the lowest operating frequency. The antenna is developed on the front of a single layer of 0.508 mm semiflexible RT/duroid 5880 substrate, with relative permittivity of 2.2, and tangent loss of 0.0009, as shown in Fig. 1. The antenna is fed using a 50 Ω CPW to provide good impedance matching. The proposed CP-antenna is designed by integrating an inverted L-stub extended from one side of the CPW ground plane in the front of the microstrip line. To improve the AR over the wideband, the width of the coplanar ground plane is considered to obtain two components of an electric field in both orthogonal x and y directions.

B. ANTENNA EVOLUTION
In this step, the design procedure is discussed to generate circular polarization by using coplanar strip (CS) based excitation technique. These technologies are characterized by easy realization, balanced circuit, and reduced crosstalk between the lines. For both asymmetric CPW and CS, closed-form expressions for the effective dielectric constant and characteristic impedance of the asymmetric CPW were derived using conformal mapping methods reported in [33].
In fact, the proposed topology underwent several development stages evolutions steps to achieve the final proposed structure as illustrated in Fig. 2. In the first step, a CPW-fed rectangular slot antenna is printed on the substrate. The length of the microstrip monopole is 13.9 mm, which is approximately a quarter wavelength at the lower cutoff frequency (referred to Ant.1). Fig. 3 shows the relevant antenna characteristics corresponding to the stages of Fig. 2. The impedance matching characteristic at stage 1 shows a resonance near 9 GHz, which is due to the quarterwavelength monopole. The AR response shows that the antenna is somewhat CP, which is attributed to the symmetrical topology of the CPW. At the second stage, a stub is added within the coplanar ground plane to the left of the transmission line to create additional electric field component along the Y-direction and generate CP radiation [34]. Moreover, to merge the CP waves at the upper and the lower bands, the dimension of the right coplanar ground plane was adjusted in they-direction. When CPW is used with a symmetric ground plane, the current directions on the edges of the ground plane are opposite to each other and, hence, cancel out in the far-field as the waves radiated by the ground plane are out of phase. As a result, the impedance bandwidth shows a clear improvement (referred to Ant.2). On the other hand, the AR response related to the stage 2 [see Fig. 3(b)] is also improved, which is due to the induction of the additional electric E-field component along L 1 , d 2 and h 2 [See Fig. 1(a)].
In addition, the antenna reflection coefficient shows a resonance at 5.93 GHz with an impedance bandwidth of 5.71-6.3 GHz in the case of Ant.2 [see Fig. 3(a)]. Also, the optimum AR frequency ≤3 dB [see ( Fig. 3(b)] is found to be wider at 6 GHz (5.86-9.05 GHz). In order to improve the S 11 and AR performances, the current stub path is extended to create inverted L-strip in the -Y direction. As a result, a further electric field, with equal magnitude and 90 o phase difference, is created to generate CP characteristic, as illustrated in Fig. 2(c) (referred to Ant.3). It is worth mentioning that using the excitation technique based on coplanar strip (CS) and asymmetric CPW, the fundamental vertical and horizontal components for the CP are generated with maintaining compact and structurally simple antenna with wide 3 dB AR bandwidth. Based on the above discussion, Table 1 propose the following design guidelines for the different design stages. To further improve the impedance matching and AR, the current path is lengthened by extending the stub to create inverted L-strip in the -Y direction.
(a)  The optimized results are shown in Fig. 3. Moreover, a wideband from 5.44 to 6.24 GHz is achieved in case of Ant.3. The simulation at this stage indicates a broadband CP characteristic from 5.27 to 8.65 GHz. The final optimized parameters are summarized in Table 2. All theoretical design phases in this study have been simulated using Computer Simulated Technology (CST) 2020 [36].   Similarly, the simulated and measured AR bandwidth of the proposed antenna, showed features of 48.56% of circular polarization axial ratio bandwidth when simulated, and 45.4% when measured. Good agreement can be observed between measurements and simulated results in terms of S 11 and AR, respectively. Regarding the circular polarization radiation mechanism, further analysis has been performed.    Fig. 7(a) depicts the proposed unit-cell geometry of the AMC plane. The designed unit-cell is based on parasitic elements' technique suggested for wideband structures [37]. This new concept consists of circular symmetric patches coupled by arms located at the center. It is realized on a semi-flexible RO5880 substrate with a thickness of H c = 0.508 mm, a relative permittivity of Ɛ r = 2.2, and loss tangent of tan δ = 0.0009. By adjusting the length of arms (d 2 ) and slots on the patch (W g ), the reflection phase bandwidth of the proposed AMC design can be easily controlled. The boundary condition of the AMC unit-cell structure is illustrated in Fig.  7(b). To simulate the reflection phase with a periodic boundary condition, each of facing walls are set to be identical to the PEC (perfect electric conductor) and the PMC (perfect magnetic conductor). The wave-port is used to simulate the reflection phase on the surface of the unit cell. The desired frequency is calculated based on equation (1) [38].

A. AMC DESIGN
Where L is the equivalent inductance and C is the capacitance associated with the periodic unit cell. From equation (1), it can be noticed that, with an increment of the inductance or the capacitance of the unit-cell, the resonance frequency will shift to the lower band. In addition, the bandwidth will decrease while increasing the capacitance. Hence, it is preferable to increase the inductance instead of the capacitance of the unit cell. In fact, the inductance can be increased without increasing the capacitance by using a thick substrate. In this case, the resonant frequency (f r ) corresponding to the effective equivalent circuit model depicted in Fig. 7(c) can be expressed as [38].
where L g and L d are the grid inductances and the dielectric slab inductance, respectively. The grid capacitance C g is the gap size between two parallel patches that can be expressed by (3) [38].
where W is the AMC unit-cell width, g is the gap between two adjacent unit-cells, and 2 a s g   , where s is the length of the unit-cell slots. Hence, the bandwidth of the AMC unitcell can be extracted using (4): shows the fabricated AMC array. The proposed design can be further miniaturized by increasing one of the following design parameters: L d , L g , C g . In fact, enhancing L d leads to a higher profile, which is not desired for wearable applications. It is noticed from equation (2) that adding slots and parasitic patches in the AMC geometry, leads to modify the current path which increases the grid inductance L g . (1) In another word, further miniaturization can be achieved by increasing the grid capacitance (C g ) which can be performed by either increasing the length of slot to gap ratio (a/g) or through dielectric loading. A suitable choice of s and g leads to an increase in the (a/g) ratio as described in (3). The inductive and capacitive effects of notches on the four corners play a prominent role in achieving broad bandwidth. Besides, the bandwidth enhancement of the AMC unit-cell design is obtained by increasing L g and decreasing C g allowed by the arm and the parasitic patch effects. In order to help figure out the impact of the equivalent circuit. This phenomenon could be observed by analyzing the surface current magnitude of the proposed unit cell at 5.8 GHz, as shown in Fig. 8. As observed, the maximum concentrated current density is mainly located along with the slot of the ring and is symmetrically distributed between the parasitic arms. This is due to the fact that by inserting the slots, the capacitive and inductive coupling enhance the AMC bandwidth. Hence, the incorporated slots can control the resonant frequency and improve the AMC unit-cell performance. In addition, as the number of slots increases, maximum current is achieved. On the other hand, the influence of the AMC design parameters is investigated to describe how the different constitutive parts affect the mechanism of the bandwidth enhancement. It can be observed from Fig. 9 that increasing s leads to shift the reflection phase (from 5.2 to 5.95 GHz). Similarly, when s is decreased, the reflection phase is also affected ( Fig. 9(a)). Moreover, by varying the radii of the big circle ring, r 1 , and the small ring, r, the reflection phase of the AMC cell changes dramatically ( Fig. 9 . On the other hand, the right-hand scale of Fig. 10(a) illustrates the high impedance feature of the AMC unit-cell, where an impedance of 16.5 KΩ is obtained. Moreover, the dispersion diagram is depicted in Fig. 10(b), where a surface wave bandgap is located between 4.2 and 6 GHz, where the AMC zero reflection frequency at 5.7 is exhibited. In the case of the integrated design, the angular stability of the AMC will influence the antenna radiation performance [39]. Therefore, this particular AMC surface was selected as a candidate for the antenna application in this work, which requires good angular stability. The reflection phase versus frequency plot for different polarization angles θ (0 o to 45 o ), under various TE and TM oblique incidence of φ=0 0 and φ=90 0 is displayed in Fig. 11. As observed, the obtained results of TE and TM polarized waves have no significant difference. It can be noticed that the wide bandwidth of the proposed AMC design is obtained in the different polarization angles of θ for various incident angles. From these results, it can be shown, from Fig. 11(a)-(b), that the proposed unit cell is highly stable as its angular margin ranges from 0 o to 45 o , thus preserving an AMC operation bandwidth of 650 MHz. It is worth mentioning that due to the symmetric design of the AMC unit cell, the results of reflection phase are similar in both polarization angles of φ=0 o and φ=90 o .

B. INTEGRATED DESIGN AND FREE SPACE RESULTS ANALYSIS
The block diagram of the far-field measurement setup and actual setup inside an anechoic chamber is illustrated in Fig.  12(a)-(b). The phase of the two field components is measured relative to the signal generator, and a rectangular horn (high linear polarization purity) serves as the source antenna. A major source of measurement errors should be eliminated, since rectangular horn antennas generate lower levels of cross-polarization over broad bands compared to circular polarized feeds [40]. The AR response, which is related to cross-polarization can be expressed by [41]: and P dB is the cross-polar power.
The AMC integrated antenna design is illustrated in Fig. 13. It is noticed that a 2×2 AMC array size was considered as a reflector to preserve the low-profile characteristic of the overall integrated design. The total footprint of the AMC array is 34.4×34.4 mm 2 , with a period of 17.2 mm. The separation between the 2×2 AMC array and the CP antenna is 3 mm, which is accounted for by means of a foam layer as shown in Fig. 13(a). Furthermore, the fabricated prototype was measured in free space and on the body in the anechoic chamber of National Institute of Scientific Research laboratory (INRS-EMT, Quebec, Canada). The simulated and the measured reflection coefficients (S 11 ) are displayed in Fig. 13(a). A good agreement is observed where the antenna system design is operating in the range of 5.45 to 6.11 GHz, whereas the measured design displayed a wideband resonance ranging from 5.27 to 6.37 GHz.

FIGURE 13. Simulated and measured results: (a) reflection coefficients, (b) realized peak gain and axial ratio.
On the other hand, the simulated and the measured realized gain for the same setup is shown in the left Y-axis scale of Fig. 13(b). It can be observed that, a high gain value is obtained within the band spectrum, due to the AMC arrangement. The AR bandwidth of the proposed antenna covers the same operating range as shown in Fig. 13(b). The simulated 3 dB AR bandwidths are approximately 20% from 5.25 to 5.89 GHz, while the measurement results indicated that the proposed antenna could achieve a bandwidth of 20.5% (5 to 6.2 GHz). Good agreement between the simulated and measured AR curves was observed. Finally, a comparison between the CP antenna without/with AMC array, in terms of realized gain and total efficiency is given in Table 3. It is noted that by adding the AMC array, the gain is increased significantly from 3.18 to 7.41 dBi at 5.8 GHz.
To validate the proposed concept, Fig. 14

A. Effects of Structural Deformation
In WBAN applications, the wearable antenna must be able to work in conformed conditions when mounted on human body surfaces. Before studying the human body loading effect, we first examine the antenna system performance under different radii of curvature values in free space. Hence, the parameter R x is used to denote the proposed antenna over cylindrical bending. As shown in Fig. 15(a), the integrated antenna with three different radii of curvature values, corresponding to 20, 40, and 60 mm along x-axis, were investigated. Bending along the y-axis is not considered for the fact that the surface current is mainly along the y-axis. Thus, bending in this direction will result in more frequency detuning as reported in [42]. A deformed CP antenna prototype is also measured to validate the accuracy of the simulations. The chosen curvatures radii are reasonable representations for different sizes of human arms and legs. Fig. 15(b) shows the simulated and the measured reflection coefficients (S 11 ) of the proposed antenna with three bending radii. We observe that the resonance frequency is maintained below -10 dB for all selected values of R x . Also, we observe that the resonance frequency is shifted 18 MHz lower when R x = 20 mm. However, when R x is increased from 40 to 60 mm, a negligible variation of S 11 is observed for both cases. In fact, it is observed that the integrated antenna is easily conformed on the cylindrical bending radius, however, the only limiting factor of the operation is the SMA connector zone, for this reason we measured only when R X = 60 mm. From these results, it can be noticed that the proposed prototype has good performances under bending scenarios, which is useful for wearable applications. The simulated and the measured radiation patterns for all antennas are plotted in Fig. 16. Due to the consistency of the simulated radiation patterns, they were verified with a single measurement of the conformal antenna. The antenna gains and efficiency comparisons for the same cases are summarized in Table 4. High gain values of the assembled antenna are achieved even in bending cases. A good agreement is observed between the numerical and experimental results. Small discrepancies were most evident in the lower half-plane axis due to the measurement's setup limitations. Nevertheless, the proposed antenna performance is shown to be robust when compared to previously reported works [26]- [32].

B. Specific Absorption Rate Analysis
Specific absorption rate (SAR) is a critical parameter for human safety when the antenna is designed to be wearable [43]. The SAR level is determined from eq. (5), where σ is the conductivity of the tissue in S/m, E is the electric field in V/m, and ρ is the mass density of the tissue in kg/m 3 .
In fact, the SAR value should comply with either the US standard of 1.6 W/kg per 1 g of tissue and below of 2 W/kg per 10 g of tissue for EU standard [44]. For off-body performance, a 100×100×33 mm 3 planar three-layered human tissue is modeled using CST microwave studio, as illustrated in Fig. 17. The model consists of the following layers; 2 mm-thick skin, 8 mm-thick fat, and 23 mm-thick muscle. This is a widely used model to analyze wearable antennas [45]. The gap distance between the antenna ground plane and the human tissue model is referred to as d. The tissues' dielectric properties of each layer of the human tissue model are listed in Table 5 [46]- [47]. The absence of a backing reflector between the antenna and the human body, can severely affect the antenna's performance [48].   Fig. 18 indicates the separation distance effects on the antenna parameters in terms of reflection coefficients and AR performances. Analyzing Fig. 18, the CP antenna preserved its resonance and AR performance at a separation of 12 mm from human body tissue. Table 6 summarizes the overall simulated gain values for various gap distances at 5.5 and 5.8 GHz, respectively. It should be mentioned that, the gain values are observed in the main lobe direction of θ=0 o . At the separation of 12 mm, simulated overall gains of 7.35 and 7.33 dBi were achieved at 5.5 and 5.8 GHz, respectively.
In addition, enhanced simulated total efficiency was observed, at 12 mm separation cases compared to other separation cases at both frequencies.
(a) (b)  The simulated SAR levels averaged over 1g on the phantom models, at 5.5 and 5.8 GHz are highlighted in Fig. 19. In this study, the calculation was based on the IEEE/IEC 62704-1 standards, with an input power of 100 mW to guarantee accurate benchmarking purposes. High values of 0.29 W/kg and 0.261 W/kg at 5.5 GHz and 5.8 GHz were achieved, respectively. Hence, the proposed antenna at a separation of 12 mm comply with the IEEE/IEC 62704-1 standards overaged over 1g and 10g of tissue. In all results presented above, the proposed antenna operates well under separation distance from the human body of 12 mm. This leads to large sizes, which are not suitable for wearable applications. Thus,   (5) incorporating a backing reflector was suggested in the form of an AMC array.

C. Human body loading
The proposed AMC array provides less backward radiation, thus, the final design should be insensitive to the human body coupling effects. To evaluate this effect, a series of experiments are performed when placed the fabricated prototype over different human body parts (chest, leg, and arm), as depicted in Fig. 20. As it can be seen, very stable reflection coefficients were maintained for all the measured cases. When placed on the chest, the proposed antenna is operating in the range from 5.37 to 5.98 GHz, where the impedance matching is still maintained in the desired band. This slight discrepancy in the S parameters is due to the highly dielectric and lossy nature of the human body at close proximity distance. To match the measurement results presented in Fig. 20(d), a full-wave electromagnetic simulation with a realistic heterogeneous (Hugo) voxel human body phantom was used to analyze the radiation performance of the proposed antenna. The heterogeneous model is composed of 32 different tissues types with a mesh size of 1×1×1 mm 3 (male, 113 kg weight, 187 cm height, 38 years old) [49]. The integrated design was placed at three different positions on the HUGO model, including the chest, arm, and leg, respectively, as illustrated in the top row of Fig. 21(a)-(c).
To reduce the simulation time, a reasonably sized portion of Hugo's body was selected for the simulation instead of simulating the proposed antenna on the entire numerical body model. In all the studied scenarios, the antenna was assumed to be separated from the body by a foam spacer with a thickness of 3 mm. After readjusting the off-body antenna orientation, the simulated radiation patterns of the proposed antenna were performed (as shown in the middle row of Fig.  21(d)-(e)). It can be seen that the proposed antenna in contact with Hugo voxel displayed a lower back-radiation for all three cases, due to the unidirectional radiation pattern type. The realized gain and the axial ratio characterizations, in free space and on-body for the same setup, are presented in the lower row of Fig. 21(f). A higher front-to-back ratio of the antenna is observed for all three cases, where the body acts as an extension on the ground plane. This increase affected the peak gain of the proposed antenna. The gain of the antenna is within the range of 6.89 to 7.84 dBi and the 3dB-AR characteristics remained unchanged from 4.9 to 6.1 GHz.
In addition, Table 7 shows a comparison of the simulated gain, FBR, and the efficiency values for all cases. It is noted that the gain values were taken in the main lobe direction of θ= 0 o . The proposed antenna exhibited high gain values when operating near to the human body model. To recall, the rationale behind the high gain values of the proposed antenna alone at separation of 12 mm is based on the fact that the human tissue acted as a reflector. This demonstrates the suitability of the integrated design for wearable applications.   Fig. 22 shows the simulated 1 g averaged SAR levels due to the proposed antenna for the three cases. As a benchmark, a power level of 100 mW accepted by the proposed antenna was chosen to evaluate the SAR performance. The maximum SAR level recorded was 0.2077 W/kg, 0.0872 W/kg, and 0.0264 W/kg, on the chest, arm, and leg, respectively. The simulated averaged SAR values over 1 g and 10 g of tissues for the three cases, are summarized in Table 8. These values are not only well below both US and EU standards, but also much smaller than those reported in the literature [18][19]- [24]- [27][28]- [51][52][53][54][55][56].
Finally, Table 9 compares the performance of the proposed design and the recent state-of-the-art antennas reported in the literature. It can be noted that the antennas [18][19][20]- [24]- [38]- [51][52][53][54][55][56] are linearly polarized (LP), which could lead to unreliable wireless links due to the polarization mismatch caused by the constant human body motion. In [26][27][28], although those antennas achieved CP radiation, they were based on rigid substrates. However, compared with the reported flexible CP antennas applied for WBAN applications, the proposed antenna has the advantages of compactness, conformability, higher gain, efficiency, wider AR bandwidth, and lowest SAR values.

V. CONCLUSION
In this paper, a compact flexible 2×2 AMC-backed planar CP monopole antenna for wearable applications has been proposed. The antenna design is based on a simple topology structure to generate a wide-band broadside CP radiation. The antenna with and without the AMC reflector enables good impedance matching in free space. Meanwhile, good performance was achieved, while loading the CP antenna at a separation distance of 12 mm from the human body model, which leads to an increase in the overall size. To minimize the separation distance and the effects of the human body on the antenna performances, a 2×2 AMC array is designed to form the antenna's backplane, and also yields toward enhancing the radiation performance. Due to using the CP approach for the antenna design, the proposed work radiator size is highly compact with a size reduction of 25% compared to recently reported works. Moreover, full-wave simulations and experimental measurements demonstrated the robustness of the assembled design for different bending conditions and human-body loading effects. The antenna system was also shown to be effective in reducing EM absorption in the human body model. With all these features, the proposed antenna is highly suitable for wearable applications.