A Multiband FSS Director Using Aperture Interdigital Structure for Wireless Communication Systems

This research paper proposes a multi-band frequency selective surface (FSS) director using the aperture interdigital technique. The FSS unit cell has been designed based on a basic bandpass filter (BPF) connected with an interdigital structure. With the proposed structure, the FSS unit cell size can be reduced from λ/2 to λ/8 caused by the slow-wave effect of interdigital capacitance loaded at the end of the transmission line in the unit cell structure. Moreover, the capacitance loaded at the end of the transmission line can control the second and the third resonance frequencies to resonate as required. The unit cell has been designed at the fundamental frequency of 1.8 GHz and the controlled second and third resonance frequencies of 3.7 GHz and 5.2 GHz, respectively. The operating frequencies of the proposed FSS unit cell cover LTE band (0-1.9 GHz), Wi-MAX band (3.22GHz-4.7GHz), and WLAN band (5.15 GHz-5.95 GHz). The size of the unit cell is very compact which is 11.53 mm × 10.77 mm. The designed unit cells are then connected as an array of 8×8, resulting in an overall size of 92.31 mm × 86.16 mm. This array has been used as a director for dipole antennas designed at the frequencies of 1.8 GHz, 2.45 GHz, 3.7 GHz, and 5.2 GHz. From the simulation, the dipole antennas with the proposed FSS director have directional gains of 4.52 dB at 1.8 GHz, 3.83 dB at 3.7 GHz, and 3.29 dB at 5.2 GHz, respectively. Measurement results show the directional gains of 3.67 dB at 1.8 GHz, 3.16 at 3.7 GHz, and 3.42 dB at 5.2 GHz, respectively. The proposed FSS director has properties of multi-band operation with a compact size that can be developed for antenna covers and radomes of multi-band wireless communications.

metamaterial reflector for dipole antennas has recently been designing [5], resulting in efficiency and gain improvement. Another method for increasing the antenna efficiency is to use an electromagnetic bandgap (EBG) structure [6][7][8]. The EBG can eliminate surface waves on the material surface, resulting in only appeared wave propagation. However, the EBG can increase the efficiency and gain of the antenna when it can only be a reflector. In addition, a technique of frequency selective surface (FSS) using a metamaterial surface has been proposed. The FSS can select a frequency range of signals to transmit or reflect. The FSS is suitable to be applied for connecting with an antenna to increase the threshold capable of transmitting antenna propagation without distorting the diffusion at a frequency range due to the coupling material. Therefore, the FSS is popularly used because it is easy to design as either a reflector or a director. It can be designed to respond to a wide range of frequencies depending on the structures. The FSSs have been studied for a wide variety of applications. The FSS associated with square patch unit cells was designed to be an antenna reflector with a structure size of approximately λ/2 and responding to only a single frequency [9]. Afterward, the FSS reflector was designed to increase the bandwidth for reflecting the antenna, enabling the antenna to use at the specified frequency, however, the size was still large [10], [11]. The FSS reflector was then designed to be compatible with wideband antennas used to enable multi-band operation but the structure was also large and very complex. Later, there was the proposed FSS reflector design with slot ground antenna using a double layer structure of ring resonators [12], [13]. The FSS has a wideband resonance which the unit cell structure of the reflector is approximately λ/2. This causes the reflector to be large and the gain of the antenna is still low. The FSSs for transmission have been also studied in a variety of formats such as characterization of a single unit cell or the FSS layer to enable multi-band frequency responses [14][15][16][17] and changing the impedance of structure to achieve multi-band frequency [18][19][20] but the design has a complex structure and the size of the unit cell is also large. From the above researches, it can be seen that the design of the FSS is as important as the design of the structure which the structure can be designed with microwave circuit design technique. It is possible to design smaller unit cells and control resonance frequencies for use in a multi-band operation [21][22][23]. This paper proposes a design of the multiband FSS director which is based on a bandpass filter (BPF) with interdigital structure, [24][25][26][27] since it has the advantage of being able to control the second resonance frequency to the desired frequency [28][29][30][31][32][33]. This unit cell is designed at the fundamental frequency of 1.8 GHz, the second resonance frequency of 3.7 GHz, and the third resonance frequency of 5.2 (LTE, Wi-MAX, and WLAN). The unit cells will be connected to act as the FSS array applied for a single band dipole antenna radiated at 1.8 GHz, 2.45 GHz, 3.7 GHz, and 5.2 GHz (one antenna one frequency). The distance between the antenna and the FSS array is approximately λ/8. The FSS array is radiated through all frequencies from the resonant antenna. This paper is divided into three sections, the first part proposes the unit cell design, the second part proposes the simulation results of the FSS unit cell connect to an array with a dipole antenna and the third part is the measurement results in the design of FSS director.

II. DESIGN OF THE UNIT CELL
This part proposes the design of the FSS unit cell, based on a multiband BPF structure connected with an interdigital section without a ground plane [31], [32]. Then, the unit cells can be arranged to form an array as FSS. When used with a radiator, the signals can transmit through the FSS with some gains. The proposed structure of the FSS unit cell is shown in Fig. 1. It can be clearly seen that the proposed unit cell is the inverted structure of the normal unit cell, in which copper and aperture are switched. It is also found that both unit cells have the same characteristics. VOLUME XX, 2017 9 The unit cell uses the step-impedance technique [26], [27] for controlling the second resonance frequency together with the interdigital technique at the end of the transmission line. Fig. 2 shows the proposed structure of the unit cell consisting of a transmission line connected to the interdigital part and also its equivalent circuit with key parameters. In this work, the unit cells are designed at the fundamental frequency (f0) of 1.8 GHz, while the controlled second (f1) and third (f2) resonance frequencies are at 3.7 GHz and 5.5 GHz, respectively applicable for LTE, Wi-MAX, and WLAN systems. From the equivalent circuit, when increasing the loaded capacitive value of the interdigital part (Ci) at the end of the transmission line, it can control the second and third resonance frequencies shifted to desired frequencies [24]. This can be effectively achieved by using the step-impedance technique, in which all important parameters can be calculated as in (1) -(3) [28].   As seen in (1), θ is the electrical length for the resonant frequency when the given values of θa0, θa1, and θa2 are electrical lengths on the transmission line at the fundamental frequency, the second and the third resonance frequencies, respectively. They can be described as the loaded capacitive of the interdigital part (Ci) at the end of the transmission line. This only has a little effect on harmonic frequency, but it has an important effect on the fundamental frequency. It is found that increasing the capacitive value causes the shifted resonant frequencies, shifting the frequency f1 more than the frequency f2. In addition, the value of Za, which is the impedance of the transmission line, also has more effect on the second resonance frequency. The transmission line using the step-impedance technique consists of parameters, in which θa is the total electrical length on the transmission line and β is the propagation constant of the transmission line. In case the second and third frequency resonance control is required, the step-impedance structure consists of two sections of the transmission lines including impedance parameters Z1 and Z2. θ0 and θ1 are the electrical lengths corresponding to Z1 and Z2, respectively. When the impedance in the transmission line Z1 larger than Z2, the K ratio will affect f1 shifted away from f0 [27][28][29][30][31][32]. Fig. 3 shows some important parameters of the proposed unit cell. The unit cell of the FSS was designed on an FR-4 print circuit which has r of 4.4 thickness of 0.035 mm, a substrate high of 1.6 mm, loss tangent of 0.025. The first dimension of this design is given in Table I. To increase or decrease capacitive load values depending on the size and number of interdigital fingers, the unit cell parameters of L5, W3, and W5 must be carefully adjusted. In addition, when increasing the loaded capacitive value, it will shift the second and the third resonant frequencies because the electrical length on the transmission line increases and is not equal to the physical length. Therefore, to make the resonant frequency shift back to the original frequency range, the size of the transmission line has to be reduced.

. The simulation configurations of the unit cell with (a) Efield on the x polarization wave (b) E-field on the y polarization wave.
Form the proposed structure, the size of the transmission line can be reduced from λ/2 to λ/8 at the fundamental frequency of 1.8 GHz. This means that using the capacitiveloaded structure can reduce the size of the unit cell. The value of Ci can be calculated by (2) when n is the number of fingers. The impedance in the transmission line can be calculated according to (3). From the above design, the unit cell has been simulated using the CST Studio Suit program to obtain the transmission characteristics. In the program, the unit cell is arranged in the z-axis and the electric field (Efield) and magnetic field (H-field) in the x-axis and the y-axis are alternately configured corresponding to the currents as shown in Fig. 4. It was found that the resonance frequency is at f0, while the harmonic frequencies of f1 and f2 can be controlled when configuring the E-field and the H-field as in Fig. 4(a). The results of frequency responses for this configuration are shown in Fig. 5(a). However, when configuring the E-field and the H-field as in Fig. 4(b), the unit cell ack as a bandpass filter on both harmonic frequencies, therefore they are unable to control for achieving desired resonance frequencies. The results of frequency responses for this configuration are shown in Fig.  5(b). This means that when applying the unit cells as FSS for the antenna system, the antenna must be placed along the xaxis so that the harmonic frequencies can be controlled.  It is found that the loaded capacitive value at the end of the transmission line must be varied by increasing or decreasing the finger and gap parameters to control the second and the third resonance frequencies of the proposed unit cell. Therefore, the interdigital part parameters including L5, W3, and W5 were studied by using CST simulation by increasing and decreasing the parameters as shown in Fig. 6. It is clearly seen that decreasing the loaded capacitive value effects the harmonic resonance frequencies of f1 and f2 shifted away from the fundamental frequency of f0. This can be noticed on the graphs that the harmonic resonance frequencies are shifted from left to right (blue and black lines) as the loaded capacitive decreased. However, increasing the loaded capacitive value will affect the harmonic resonance frequencies of f1 and f2 moved toward the fundamental resonance frequency of f0. The results will be applied to the final design of the proposed unit cell.  The simulation results of the unit cell using interdigital structure (a) insertion losses (S21) and (b) return losses (S11).
In our design, the second and third resonance frequencies are moved closer to the fundamental frequency of f0 as a result of capacitive value increment, therefore the electrical length on the transmission line is not equal to the physical length. If the fundamental resonance frequency is designed to resonate at the original frequency, the length of the transmission line must be reduced to lesser than half its original wavelength. It is found that the length of the transmission line can be reduced from λ/2 to λ/8. In addition, to design the unit cell having the second and the third resonance frequencies to be able to resonate at the desired frequencies, the step-impedance technique on the transmission line is used together with the interdigital structure. By using the step-impedance technique, the harmonic frequencies can be further controlled. Therefore, the key parameters of the step-impedance structure of W4 and W6 were studied using the CST simulation as the result shown in Fig. 7.  Form Fig. 7, it is found that when using technique stepimpedance with the ratio between Z1 and Z2 or ratio of K > 1, it will cause frequencies of f1 and f2 to shift away from the frequency of f0, which has a great effect in shifting the second resonance frequency as shown in the red line. If Z1 and Z2 or ratio of K< 1, it will cause the frequencies of f1 and f2 to shift closer to the frequency of f0 as shown in the black line. In our design, the suitable impedances for producing the desired resonance frequencies include Z1 = 100 ohms and Z2 = 133 ohms. After designing the unit cell, the E-field and Hfield values on the unit cell structure were studied as shown in Fig. 8. It is found that at the resonance frequency f0 the Efield is dense on the transmission line and the interdigital interval. At the harmonic resonance frequency of f1, it is found that the E-field density is observed on the stepimpedance transmission line and interdigital part. At the harmonic resonance frequency of f2, it has a density of the Efield as same as the resonant frequency of f0. The results from the simulation are consistent with the equation and simulation results as mentioned above. The H-field density on the structure of the unit cell was studied by CST simulation as the results shown in Fig. 9. It is found that at all resonant frequencies, the transmission line and the interdigital intervals have fewer H-field densities compared with the E-field densities due to the arrangement of the unit cell structure. Then the current distribution on the unit cell was simulated as the results shown in Fig. 10. It is found that at resonant frequencies of f0 and f2, there are high current distribution in the transmission line and interdigital part. In the resonant frequency of f1, there is more current density on the stepimpedance transmission line and interdigital part. The simulation results consistent with the design equations and directly correspond to the E-fields. When comparing the size of the proposed unit cell with the conventional metallic patch unit cell in Fig. 11, it is found that the size of the proposed unit cell is significantly smaller because of using the design techniques mentioned above. The approximate size of the proposed unit cell structure is λ/8, while the conventional structure is approximately λ/2. In addition, the proposed unit cell structure can be operated for three controlled resonance  frequencies, besides the conventional metallic patch unit cell structure can only be operated for a fundamental frequency and the uncontrollable second and third resonant frequencies. Therefore, the proposed unit cell can be easily applied for a wide variety of multiband antenna applications.
In our design, the unit cell was intentionally designed to operate at the fundamental frequency of 1.8 GHz with the lowpass transmission response and the bandwidth of 1.9 GHz (0-1.9GHz). Also, it has the second resonance frequency of 3.7 GHz with the bandpass transmission response and the bandwidth of 1.48 GHz (3.22GHz-4.7GHz), and the third resonance frequency of 5.2 GHz with the bandpass transmission response and the bandwidth of 0.8 GHz (5.15GHz-5.95GHz) as shown in Fig. 12. The insertion losses at the resonance frequencies are 2.1 dB, 1.36 dB and1.21dB, respectively, and also the return losses are -7.4 dB, -23 dB and -15 dB, respectively. When comparing phase angle responses, they consistent with phase reversals for the non-transmitted phase corresponding to the return losses at all resonance frequencies. This will allow all three frequencies to transmit through the controlled frequency bands, resulting in resonance covering the LTE, Wi-MAX, and WLAN systems.

III. DESIGN OF FSS DIRECTOR WITH THE DIPOLE ANTENNA
After designing the unit cell in the previous section, the proposed unit cells are connected to form an array as frequency selective surface (FSS). The number of unit cells in this array is 8×8 unit cells corresponding to the size of the dipole antenna as shown in Fig. 13(a). The overall size of the designed FSS array is 92.32 mm × 86.16 mm. Fig. 13(b) shows the simulation results of the FSS array structure and Fig. 13(c) shows the simulation gain of FSS array at 1-7 GHz. It is found that the FSS can transmit the signals over the designed frequency ranges as same as the single unit cell. However, the insertion loss and return loss responses may have little differences caused by the effects from the unit cell connection. Next, a dipole antenna is designed as shown in Fig. 14 and it will be placed over the FSS array. The dipole antenna dimensions for three operating frequencies are listed in Table II. The dipole antennas were designed with copper metal tubes at each operating frequency of 1.8 GHz, 3.7 GHz, and 5.2 GHz. A low-loss cable and a balun were connected to the dipole antenna. From the simulation results, each dipole antenna has a return loss (S11) lesser than -10 dB at the resonance frequencies and gains of the antenna are about 2 dB, 1.97 dB, and 1.86 dB, respectively. Then, the proposed FSS was employed to be a director for each dipole antenna VOLUME XX, 2017 9 by placing it behind the dipole as shown in Fig. 15(a). The boundary conditions around the dipole antenna and the FSS array are set as shown in Fig. 15(b). The distance between the dipole antenna and the FSS array at all resonance frequencies is about /8, which is very small comparing with the conventional antenna structure. The simulation results of the dipole antenna with an FSS array at all resonance frequencies are shown in Fig. 16. The is found that the return losses (S11) lesser than -10 dB at 1.

IV. MEASUREMENT RESULTS
This section proposes the implementation and testing of the dipole antenna with the proposed FSS director. The FSS director was built on the FR-4 printed circuit board and by using a milling machine process. Dipole antennas were made of copper tubes connected by a low-loss signal cable and an SMA-51 type connector, and a balun to make it a balanced line. Three dipole antennas were constructed for all three resonant frequencies. Fig. 18(a) shows pictures of the fabricated FSS director and the dipole antenna. The antenna system including the dipole antenna and the FSS director were tested in the chamber room as shown in Figure 18. Measured results show that at the three resonant frequencies, the antennas have return losses (S11) lesser than -10 dB as shown in Fig. 19

V. CONCLUSION
The proposed unit cell is based on the bandpass filter (BPF) structure using aperture interdigital and step-impedance techniques. It can be used for three operating frequency bands, which all bands can be controlled as desired by adjusting the loaded capacitive value at the end of the transmission line. Also, the size of the proposed unit cell is very compact due to the length of the transmission line in the unit cell structure is reduced from /2 to /8 caused by the slow-wave effect. The proposed unit cells have been used to design the FSS director for the dipole antennas to increase the gains. Also, with the proposed FSS director, the distance between the FSS and the dipole antenna is about /8, which is very small compared with the conventional antenna system. The FSS director has the best performance at both Eplane and H-plane in co polarization. The simulation results of dipole antenna with the proposed FSS director at all of frequencies have return loss lesser than -10 dB and the gains in E-plane co-polarization of 4.52 dB at 1.8 GHz, 3.83 dB at 3.7 GHz, and 3.29 dB at 5.2 GHz. In the measured results, the dipole antennas with the proposed FSS director at all resonance frequencies have the return losses lesser than -10 dB and the gains in E-plane co-polarization of 3.67 dB at 1.8 GHz, 3.16 dB at 3.7 GHz, and 3.42 dB at 5.2 GHz, respectively. For H-plane co-polarization, the antenna gains are 3.53 dB at 1.8 GHz, 3.18 dB at 3.7 GHz, and 3.22 dB at 5.2 GHz, respectively. From the design of multiband FSS it can be seen that the simulation results and the measurement results are similar. The proposed FSS director can operate covering LTE, Wi-MAX, and WLAN systems.. However, the proposed FSS director can be also applied for any multiband wireless communication system such as space technology, especially it can be used as a part of cover or radome of antenna system for higher gain and compact size.