Wideband SIW Half-Mode/Quarter-Mode-Fed Microstrip Patch Complementary Antennas With Back Radiation Suppression

Complementary sources concept and half-mode/quarter-mode substrate integrated waveguide (HMSIW/QMSIW) technology are introduced in this paper to improve the back-radiation suppression and bandwidth of microstrip patch antennas (MPAs). The HMSIW cavity as the feeding structure brings an additional resonant frequency to the inherent resonant mode of typical MPAs, which efficiently extends the bandwidth of the MPAs. In addition, while stimulating the MPA mode, the HMSIW cavity drives a monopole by slightly breaking the shorting wall of the HMSIW cavity and loading a printed monopole. The loaded monopole and the equivalent magnetic currents of the MPA construct an orthogonal complementary source model that indicates back radiation suppression. A differential feeding scheme is also employed to enhance the radiation performance. The dual-polarized model, achieved by replacing the HMSIW cavity with QMSIW cavities, is also addressed and even fabricated and tested. The measured results show a differential impedance bandwidth of 15.4% from 1.91 GHz to 2.23 GHz with a differential port-to-port isolation of 38 dB. In the operating band, the average gain is 7.5 dBi, and the front-to-back ratio (FBR) is over 18 dB with a maximum value of 34 dB.


I. INTRODUCTION
DUE to their attractive properties, including low cost, light weight, and ease of integration, microstrip patch antennas (MPAs) have been widely adopted in many modern communication systems [1]. Because of this, MPAs have been thoroughly studied and developed since the 1950s. Traditional MPAs always suffer from narrow bandwidths [2], [3]. Since the bandwidth can be improved by increasing the thickness of the substrate, this is not a qualitative change.
The associate editor coordinating the review of this manuscript and approving it for publication was Tutku Karacolak .
All of these techniques can enhance the bandwidth of MPAs to meet the bandwidth requirements of many communication systems but are accompanied by the deterioration of other types of antenna performance, such as high cross-polarization [8], [9], large side lobes [14]- [17] or strong back radiation [12], [13], [18].
Although these broadband technologies cause more or less performance degradation, almost all the types of degradation have related specialized research and literature related to overcoming them, including cross-polarization suppression [19], [20], side-lobe suppression [14], harmonic suppression [21]- [23] and so on. However, studies and analyses on back radiation suppression are relatively rare. The studies focusing on improving the front-to-back ratio (FBR) of MPAs always utilize metamaterial reflectors [24], [25], which usually require additional substrate layers or a large size. Indeed, as communication systems become increasingly compact, the aperture that can be supplied to the antenna decreases.
It is also increasingly unacceptable to utilize a sufficiently large ground plane to achieve a high FBR for MPAs. Implementing a broadband microstrip antenna in a compact space while suppressing the backward radiation of the antenna has become a problem worthy of attention.
In this paper, we introduce half-mode/quarter-mode substrate integrated waveguide (HMSIW/QMSIW) techniques [26], [27] to excite MPAs for enhanced bandwidth. Meanwhile, complementary sources technology [28], [29] is also adopted to improve the FBR performance. When the MPAs are stimulated, a loaded monopole is also fed by the improved HMSIW/QMSIW cavity, so orthogonal complementary sources are constructed to suppress the back radiation [30]- [32]. In addition, a differential feeding scheme is also adopted. On other hand, differential feeding scheme can ensure a symmetrical radiation pattern and low cross-polarization [14], [15], [33]. On the other hand, integrating differential antennas with differential circuits can achieve a higher integration and efficiency [34], [35]. A differential dual-polarized MPA, achieved by two pairs of orthogonal monopole-loaded QMSIW cavities, is fabricated and tested. The measured results show a bandwidth enhanced by 15.4% with a high differential port-to-port isolation of over 38 dB. Due to the introduction of complementary source technology, the maximum value of FBR is extended to 34 dB with an average gain of 7.5 dBi.

II. PRINCIPLE OF OPERATION
A. HMSIW/QMSIW CONCEPT A typical rectangular SIW resonator under the dominant mode is shown in Fig. 1, and its resonant characteristics have been thoroughly analyzed in previous reports [26]. Due to the high ratio of width to height for the SIW resonator, the electric-field distribution is nearly uniform in the z-direction, so the magnetic field is always vertical to the symmetrical planes (AA', BB', CC', DD'), and these symmetric planes of the SIW are equivalent to magnetic walls. Hence, when half of the metal cover along the symmetry plane ( Fig. 1(b) and Fig. 1(d)), or even three quarters ( Fig. 1(c) and Fig. 1(e)) is removed, the electromagnetic field distribution in the rest can remain approximately intact.
Although in the HMSIW/QMSIW cavity ( Fig. 1(c) and Fig. 1(e)), the field distribution can still be kept roughly the same as that in the original SIW cavity ( Fig. 1(a)), the electromagnetic energy is inevitably radiated in part from the open edges, and this characteristic is even used as a radiation source in antenna design in the millimeter wave (mmW) band [26], [27]. The radiation process can be equivalent to magnetic current radiation sources parallel to the open edge because the edge electric field in the HMSIW/QMSIW cavity is always vertical to the open edge. Hence, the magnetic field distribution is parallel to the open edges of the HMSIW/QMSIW cavity, which makes it possible to work as a magnetically excited source in MPAs.

B. HMSIW-EXCITED MPA
The HMSIW/QMSIW cavities can work as antennas in the mmW band, while they are not efficient enough in the lower frequency band due to the ultralow profile. However, HMSIW cavities can be applied in magnetic-excited MPAs working as a feeding structure (Fig. 2) as an option that can be used instead of aperture-coupling feeding [12], [13] and torsion-coil coupling feeding [36]. Taking the HMSIW model shown in Fig. 1(b) as an example, the equivalent magnetic currents generated by the HMSIW cavity can also excite MPAs, as shown in Fig. 2(b)). Obviously, this feeding scheme can avoid a complicated 3D structure and even keep the ground plane integrated.  [12], [13], [36] and (b) HMSIW-fed MPA model.
The typical MPA has an inherent resonant mode, and introducing the HMSIW cavity as the feeding structure can introduce an additional resonance mode. The dual-mode resonance can efficiently expand the bandwidth compared to the original single-mode resonance of the typical MPA. The equivalent circuit of HMSIW-fed MPA is shown in Fig. 3(a), and dual-mode resonance feature of the two-order resonator ( Fig. 3(b)) can be noticed and that is the basis of broadband performance. Obviously, the HMSIW cavity works only as the excitation of MPAs and can be regarded as a lossless resonator if ignoring the dielectric loss and metal loss, so the radiation mode throughout the wide band is a fixed MPA mode, which indicates a stable radiation performance, including a stable gain and radiation pattern.

C. SIW-DRIVEN MONOPOLE
As shown in Fig. 2(b), the radiating magnetic currents are parallel to the edges of the radiation patch. Hence, to construct the complementary sources for back radiation suppression, it is necessary to construct an electric antenna that is placed perpendicular to the magnetic current element. Dipoles and monopoles are common electrical antenna options, especially bow-tie-shaped dipoles and triangular monopoles, which are always able to achieve excellent wide bandwidth performance.
Therefore, a new model, shown in Fig. 4(a), is considered, and several shorting vias of the SIW cavity are cancelled where a trapezoidal stub is loaded simultaneously. The minor damage does not substantially alter the characteristics of the SIW cavity, so it can maintain its dominant mode state. The junction can also maintain the voltage zero state (wave node), and the end of the monopole is the voltage antinode, so the loaded stub can work in a monopole state. The field distribution feature in HMSIW is similar to that of the original SIW cavity. The same operation can operate in the HMSIW cavity, and the monopole-integrated HMSIW model is shown in Fig. 4(b). Then, using the improved HMSIW cavity to replace the original HMSIW-fed MPAs (Fig. 4(c)), a complementary-source-based MPA is achieved. As shown in Fig. 5(a), the magnetic current is orthogonal to the electric   sources, so the foundation of enhanced back radiation suppression is available ( Fig. 5(b)).

III. ANTENNA ANALYSIS A. DIFFERENTIAL HMSIW-FED COMPLEMENTARY MPAs
Due to the inherent high pattern symmetry and low cross-polarization performance of the differential antenna, a differential feeding scheme is introduced in these HMSIW-fed MPAs for more obvious analysis of the back radiation suppression characteristics. The configuration of the differential HMSIW-fed complementary MPA is shown in Fig. 6. The radiation patch of the MPAs is the same size as the ground plane. Two differentially fed HMSIW cavities simultaneously drive the MPA and a pair of monopoles.
To better reveal the improvement from the differential HMSIW-fed complementary concept, two reference antennas (Ant. I and Ant. II) are introduced here for comparison with the proposed complementary MPA (Ant. III). As shown in Fig. 7(a), Ant. I is a typical MPA driven by a pair differential probe, and Ant. II is a conventional differential HMSIW-fed MPA without complementary sources. For a fair VOLUME 9, 2021 comparison, the ground planes of both Ant. I and Ant. II are expanded to ensure that the three models have the same size.
As shown in Fig. 7(a), the differential-mode reflection coefficient S dd11 and FBR curves are compared. The resonant characteristics change from single-mode resonance to dual-mode resonance when the model evolved from Ant. I to Ant. II, which indicates an enhanced impedance bandwidth. A comparison of Ant. II and Ant. III shows that the shrinking ground-plane size and the additional monopoles have no essential influence on the port performance. The introduced complementary sources concept substantially improves the FBR performance of the MPA. The maximum boost is at 2.1 GHz over 30 dB, which is a 16 dB improvement over that of Ant. II. The FBR curves of Ant. I and Ant. II exhibit similar trends but are worse overall than those of Ant. I. This occurs because to make the three models resonate at approximately the same frequency, the radiation patch of Ant. I is larger than those of the other two. From another perspective, as compared in Fig. pattern, the radiation patterns of two reference antennas (Ant. I and Ant. II) exhibit an obvious back lobe, while the proposed differential HMSIW-fed complementary MPA (Ant. III) exhibits obvious zero radiation in the back direction. This radiation pattern comparison also demonstrates the ability of the introduced complementary source technology to suppress the backward radiation of a compact microstrip antenna.

B. DIFFERENTIAL QMSIW-FED DUAL-POLARIZED COMPLEMENTARY MPAs
Compared to an HMSIW resonant structure, the QMSIW resonant cavity has significant advantages and a more compact structure that make dual-polarized complementary MPAs possible. As shown in Fig. 8, two pairs of orthogonally placed differential QMSIW cavities excite the orthogonal TM 10 and TM 01 modes of MPAs. Naturally, every QMISW cavity is loaded with a monopole, and the whole structure is a rotationally symmetric structure, which conceals a theoretically infinite differential port-to-port isolation. Note that the HMSIW cavity is designed by selecting coordinate axis segmentation ( Fig. 1(b)), while the QMSIW cavity is based on diagonal segmentation ( Fig. 1(e)), which can be attributed to a complete shorting edge of the SIW cavity and facilitates the integration of monopoles.  There is a detailed design in which a tiny slot is added on the top layer of the QMSIW cavity, which is considered for turning the impedance match. In Fig. 9, the effect of the length of the turning slot is demonstrated. The wide frequency band divides into two isolated bands without the turning slot. Introducing a turning slot with increasing length has no essential effect on the lower resonant frequency, while the higher resonant point is sensitive to the variation. The slot can make the surface current path of the QMSIW cavity roundabout, which can cause the resonant frequency of the QMSIW cavity to shift to the low end, like the trend of the higher resonant point in Fig. 9. While a long slot will damage the coupling mechanism between the QMSIW cavity and the MPA, as shown in Fig. 9, the port performance will dramatically degrade when the turning slot is too long.
Since the loaded monopole can essentially change the radiation characteristics of the MPA, the monopole is worth focusing on. First, the triangular monopole is a typical wideband radiator that can simplify the optimization work and make it easier to adjust the two components of the complementary source with similar radiation amplitudes. Second, the length of the monopole is the key parameter for the FBR, as shown in Fig. 10, and changing the length of the monopole changes the maximum value of the FBR and its corresponding frequency. The length of the monopole can affect the radiation intensity of the monopole, which is the magnitude of the 8-shaped pattern in Fig. 5(b). Theoretically, a suitable monopole radiation intensity based on a decent monopole length can realize perfect backward radiation suppression. The monopoles also have an effect on the port performance, mainly at higher frequency resonance points, while the port characteristics are generally a good match. Hence, the loaded monopoles mainly influence the radiation performance without having an essential effect on the port performance of the MPA.

IV. EXPERIMENT AND DISCUSSION
To verify the previous analysis and the effectiveness of the improved HMSIW/QMSIW-cavity feeding mechanism, as shown in Fig. 11, a differential dual-polarized prototype based on Fig. 8 is fabricated and tested. The whole model  is constructed with an aluminum patch as the radiator and a substrate designed on the QMSIW cavities and loaded monopoles.

A. PORT PERFORMANCE
The measured reflection coefficient curves and differential port-to-port isolation are compared with simulated ones in Fig. 12 and Fig. 13, respectively. The simulated |S dd11 | and |S dd22 | are perfectly coincident, so they are represented by a curve. The good agreement between the measured |S dd11 | and |S dd22 | also verifies this feature. Based on the standard of |S dd11 |/|S dd22 | < −10 dB, the measured impedance bandwidth (15.4%, 1.91 GHz -2.23 GHz) is slightly narrower than the simulated results (16.3%, 1.92 GHz -2.26 GHz). The main difference is that the higher resonant frequency shift to the low end. Based on the previous analysis (Section III, B), the higher resonant point is mainly determined by the QMSIW cavities, so the measured result error can be attributed to the potential thickness and dielectric constant instability of the dielectric substrate. There is an VOLUME 9, 2021  obvious gap between the simulated differential port-to-port isolation and the measured results caused by fabrication error. At the same time, the test error also makes the measured isolation curve not as smooth as the simulation one.
The reflection coefficients and differential isolations are defined in [37]. |S dd11 |, |S dd22 |, |S dd12 | and |S dd21 | can be transformed from the single-end S-parameters of the four-port network using the following formulas.

B. RADIATION PERFORMANCE
Comparing the measured gain and simulated gain in Fig. 14 shows good agreement between the measured results and simulated results. The stable antenna gain in the operating band is approximately 7.5 dBi, and the fluctuation in the antenna gain is lower than 1 dB. Compared to the simulated results, the measured 1-dB gain bandwidth is somewhat narrow with the same trend as the reflection coefficient curves. In addition, the antenna has an excellent antenna efficiency over 0.95 in the operating band (Fig. 14).
The FBR performance is the main discussion point of this paper. The simulated results show a greater than 18 dB FBR over the whole band with a maximum value of 38 dB at  2.04 GHz, while this parameter is greater than 18 dB in the measured band with a maximum value of 34 dB at 2.05 GHz. In general, the simulation results and measured results exhibit the same trends and acceptable agreement.
The normalized radiation patterns in the E-plane/H-plane at 1.94 GHz, 2.06 GHz and 2.2 GHz of the fabricated prototype when differential port_1 is excited are compared in Fig. 16. Similar radiation patterns can be observed when differential port_2 is driven, but these results are not shown here for simplicity. Great agreement between the measured copolarization and the simulated results can be found. The measured cross-polarization is approximately 20 dB, while the simulated cross-polarization is not described because it is less than the range shown in Fig. 16. The large gap between the simulated cross-polarization and experiment is caused by assembly errors and the limited dynamic range of the test system. The excellent back radiation suppression performance can also be confirmed by the measured radiation pattern. In particular, Fig. 16 (b) shows a clear and precise back-radiation zero.

C. DISCUSSION
Interestingly, the radiation pattern of the complementary antennas should have been heart-type, as shown in Fig. 5 (b), while both the simulated and measured results, as shown in Fig. 16, show that this is the case only for the radiation pattern in the H-plane and that the radiation pattern in the E-plane is clearly different. There are two radiation quasizeros at θ = −90 • and θ = 90 • . The reason can be traced back to the radiation source model. In the E-plane, the radiation sources can be considered to be a two-element  array with complementary units, and the distance between the two complementary elements is approximately the width of the radiation patch, which is approximately 0.4λ 0 in these proposed MPAs. Therefore, the theoretical radiation pattern should be the typical complementary source radiation pattern synthesized with an equivalent amplitude and phase excitation binary array factor with a unit space of 0.4λ 0 , as illustrated in Fig. 17. Obviously, acceptable agreement can be found in a comparison of the synthetic theoretical radiation pattern in Fig. 17 and the simulated/measured radiation pattern in Fig. 16 (b).
Finally, a comparison between this prototype and previously reported differential wideband MPAs is listed in Table 1. Obviously, this work developed a decent wideband performance. More importantly, this prototype achieved an excellent FBR performance based on a compact size.

V. CONCLUSION
A new wideband MPA is proposed in this paper. The impedance bandwidth of the proposed antenna is enhanced by the HMSIW/QMSIW cavity as the feeding structure. Additionally, the concept of complementary radiation sources is applied to suppress the back radiation of the MPAs. The orthogonal complementary sources are constructed by the equivalent radiation magnetic currents of the MPAs and a loaded monopole fed by the HMSIW/QMSIW cavity. To highlight the effect of the introduction of complementary sources, a differential feeding scheme is adopted for symmetrical radiation performance. Both the line-polarized model differentially fed by the improved HMSIW cavity and the dual-polarized model differentially fed by the improved QMSIW cavity are analyzed, and the dual-polarized case is fabricated and tested. The measured results show an enhanced bandwidth of 15.4% with a high differential port-to-port isolation of over 38 dB and a maximum FBR of 34 dB with an average gain of 7.5 dBi in the operating band.