An X/Ku Dual-Band Switch-Free Reconfigurable GaAs LNA MMIC Based on Coupled Line

This article presents an X/Ku dual-band switch-free reconfigurable GaAs low-noise amplifier (LNA) realized by inter-stage and output-stage coupled lines. This article is the first switch-free reconfigurable LNA design in the coupled lines structure. After amplified by the broadband drive stage, the input signal is divided into two parallel single-band stages (consists of a high-band stage and a low-band stage) by the proposed inter-stage coupled line. Two split-band signals are combined by the proposed output-stage coupled line into the output port after amplified by the single-band stages. The proposed coupled lines are also included in LNA matching networks. Dual-band operation of the proposed reconfigurable LNA is achieved by turning off drain voltages of the unused single-band transistors. The reconfigurable LNA is designed in a 0.15-<inline-formula> <tex-math notation="LaTeX">$\mu \text{m}$ </tex-math></inline-formula> E-mode GaAs pHEMT process. The fabricated LNA features small-signal gains of 25-25.2/20.1-28 dB, noise figure (NF) of 1.28-1.41/1.23-1.51 dB, and output 1-dB compression point of 0.5-1.7/2.2-5 dBm over 8-10/12-20 GHz, while consuming 76/91 mA of dc current, with a size of <inline-formula> <tex-math notation="LaTeX">$2.0\times 1.8$ </tex-math></inline-formula> mm<sup>2</sup>.


I. INTRODUCTION
As one of the most critical active components, low noise amplifiers (LNAs) are highly demanded in modern active phased array systems such as multi-standard and multi-band transceivers [1]. To expand the dynamic range and improve the sensitivity of a receiver system, we expect to achieve low noise figure (NF), reasonable gain, high linearity as well as low power consumption for LNAs, which further introduces challenges on LNAs design [2]. In recent decades, extensive researches have been carried out on wideband LNAs. Distributed amplifier (DA) topology is most promising to be applied in an ultra-wideband LNA design [3]- [8]. Unfortunately, DA is not preferable in many situations for its relatively high NF, large chip size, and low unity gain [2]. Many other bandwidth enhancement techniques have also been presented in [9]- [15]. These proposed techniques can extend the bandwidth but also lead to the degradation of other RF performance [13]. Furthermore, the sensitivity of The associate editor coordinating the review of this manuscript and approving it for publication was Vittorio Camarchia . a receiver system with wideband LNAs may degrade significantly because the wideband LNAs amplify the desired signals as well as unwanted signals simultaneously.
Another approach to developing wideband receiver systems with good interference rejection is using reconfigurable LNAs. Many multi-band amplifiers configurations have been presented in the literature [16]- [22]. In [16], a 1.9-2.4 GHz reconfigurable LNA with a continuous tunable input matching network is proposed by utilizing tunable inductors. [17] reports a 2.4-5.4 GHz reconfigurable LNA with a broadband input stage and a band-selective stage using a multi-tap switching inductor and varactors. A reconfigurable multimode LNA employing a switched multi-tap transformer in the input matching network has been proposed in [18]. However, the tunable inductors, varactors, and RF switches mentioned above usually need extra power supplies and cannot be easily implemented at high frequencies for its associated loss. A 6-18 GHz switchless dual-band, dual-mode power amplifier (PA) has been reported in [19] using a coupled-line-based diplexer, which is similar to the coupled lines used in this article.
In this article, an X/Ku dual-band switch-free reconfigurable LNA is presented. The reconfigurable feature is realized by the inter-stage and output-stage coupled lines, which are also included in the LNA matching networks. The proposed reconfigurable LNA has been fabricated using WIN 0.15-µm enhancement-mode (E-mode) GaAs process. The die area of the proposed MMIC LNA is 2.0 × 1.8 mm 2 , including all RF and DC power pads. With the frequency reconfigurable capability, the LNA exhibits 25-25.2/20.1-28 dB small-signal gain, 1.28-1.41/1.23-1.5 dB NF, and output 1-dB compression point of 0.5-1.7/2.2-5 dBm over  This article is organized as follows. Section II introduces the theoretical analysis and reconfigurable mechanism of the proposed microstrip coupled lines. Section III presents the detailed circuit design of the proposed reconfigurable LNA. Measurement results of the proposed reconfigurable LNA are presented in Section IV, followed by a conclusion in Section V.

II. THEORETICAL ANALYSIS OF THE PROPOSED COUPLED LINES
The reconfigurable feature of the proposed dual-band LNA is realized by the inter-stage and output-stage coupled lines. To better understand the mechanism of the dual-band operation of the proposed coupled lines, an analysis of a conventional coupled microstrip line is introduced. Such a line is shown in Fig. 1(a).
In a conventional coupled line structure, the load impedance Z load is assumed to be identical to the characteristic impedance Z 0 of the coupled-line. In this case, the output voltage of the coupled port (port 3) and through port (port 2) as a function of the input voltage V in can be given as [23]: Voltage coupling factor of port 3 as a function of the coupling coefficient, K and electrical length, θ can be given by: K is the coupling coefficient of the coupled line, and θ is the electrical length of the coupled line. From (1) and (2), we can see that at very low frequencies or very short line length (θ π/2), virtually all input power is transmitted to port 2, with none being coupled to port 3. For θ = π/2, the first maximum coupled power transmitted to port 3 is  shown in Fig. 2. For K = 0.7, we can see that the maximum coupled power is approximately -3 dB with a long-coupled line (θ = π/2). For this reason, the conventional coupled line structure in Fig. 1(a) may not be applicable in the proposed LNA.
A modified structure of the coupled line is shown in Fig. 1(b). The isolated port (port 4) is grounded. For ideal VOLUME 8, 2020 high-band frequencies operation, the shunt capacitance C 1 at the through port (port 2) performs a short circuit. For the low-band frequencies operation, the shunt capacitance C 1 is absorbed in the input impedance matching network of the low-band LNA. The output signal at the through port is reflected with a 180 • phase shift and enters the coupled line again. Then the signal is coupled to the isolated port and reflected with a 180 • phase shift and comes out of the coupled port (port 3) with the original coupled signal eventually. The output voltage of isolated port (port 4) and coupled port (port 3) as a function of the input voltage V in can be given by: Coupled power of port 3 as a function of the coupling coefficient, K and electrical length, θ can be given by: In (5), we can see that the eventually coupled signal coming out of the coupled port consists of two parts: C ·V in and C ·T 2 · V in . The phase difference between the two parts is identical to the double electrical length of the coupled line. As shown in Fig. 3, when the electrical length of the coupled line is λ/4, the phase difference between two parts of the coupled power is 180 • , then the minimum coupled power is obtained. Fig. 3 illustrates the coupled power in port 3 of the coupled line shown in Fig. 1(b) according to K , coupling coefficient of the coupled line. For a definite K , a maximum coupled power can be obtained when the coupled line has a 30 • ∼45 • or 135 • ∼150 • electrical length. When K = 0.7, the maximum coupled power can be obtained is approximately -4 dB and is not sufficient to be used in the inter-stage of the LNA. The reason for the insufficient coupled power lies in the phase difference between C · V in and C · T 2 · V in , the two parts of the coupled power. To tackle this problem, we added a shunt capacitance C 2 to the isolated port to make up for the phase difference between two parts of the coupled power.
Another possible inter-stage coupled line structure is shown in Fig. 1(c). In this structure, the coupled port (port 3) is grounded, and coupled power comes out of the isolated port. The output voltage of port 3 and port 4 as a function of the input voltage V in can be given by: For ideal high-band frequencies operation, the through port (port 2) and coupled port (port 3) perform short circuits. The output signal at the through port is reflected with a 180 • phase shift, enters the coupled line, and comes out of the isolated port (port 4). The coupled signal at the coupled port is reflected with a 180 • phase shift, and comes out of the isolated port finally. From (8), we can see that the eventual signal at the isolated port consists of two parts: output signal at the through port and coupled signal at the coupled port. Moreover, the two parts are in phase.
The output coupled power of port 4 as a function of the coupling coefficient, K and electrical length, θ is given by: Specially, for θ = 90 • , coupled power can be given by:    Fig. 1(c) according to different coupling coefficient, K and electrical length, θ of the coupled line. From Fig. 4(a), for a constant θ, we can see that the coupled power increases to the peak and then decreases as K increases. When θ rises, the value of K needed to achieve the maximum coupled power is smaller. For K is 0.58, the maximum coupled power can be obtained is approximately -0.35 dB when θ is 90 • . Fig. 5(a) is a circuit schematic of the proposed coupled line B. The length of the coupled line is 600 µm with 30 • ∼ 40 • electrical length over 12-18 GHz. Width and space are 7 µm and 5 µm with K, the coupling coefficient is 0.58, approximately. A shunt capacitance of C 2 = 173 fF is added to the isolated port to increase coupled power. Fig. 5(b) shows the simulation results of the proposed coupled line. Over 14-20 GHz, simulation results show that coupled power is approximately -2 dB, which is about 3 dB higher than analytical results shown in Fig. 3 due to the elimination of the phase difference between the two coupled power paths. Thus, the proposed coupled line shown in Fig. 5 can be applied in the inter-stage of the reconfigurable LNA. Fig. 6(a) is a circuit schematic of the proposed coupled line C in Fig. 1(c). Length, width, and space between coupled lines in Fig. 1(b) and (c) are identical. A shunt capacitance of C 3 = 0.3 pF is added to the isolated port to increase coupled power. Fig. 6(b) shows the simulation results of the proposed coupled line C. Simulation results show that coupled power is approximately -1.8 dB over 12-18 GHz. It can be found that the coupled power of the two coupled lines shown  in Fig. 1(b) and (c) is comparable. Considering the convenience of layout, coupled line B showed in Fig. 1(b) is finally chosen as the inter-stage coupled line.
Schematic and simulation results of the output-stage coupled line are presented in Fig. 7(a) and (b). It's a symmetric structure of the inter-stage coupled line B with optimized C 2 and width of the coupled line. Insertion loss is less than 2 dB over 8-12 /14-20 GHz.

III. CIRCUIT DESIGN
The topology of the proposed dual-band switch-free reconfigurable LNA is shown in Fig. 8. The LNA consists of a broadband drive stage, single-band stages (high-band stage and low-band stage), and inter-stage/output-stage coupled-lines. All transistors in the proposed LNA are designed in commonsource (CS) architectures. Inductive source degeneration is used to achieve simultaneous NF and impedance matching [24]. Moreover, source inductance can help improve the stability of a transistor. The inductances are implemented via long bent transmission lines to achieve a compact chip area at the cost of a little more insertion loss. Instead of quarterwave long transformers which are usually used as bias circuits to block RF signal, narrow long bent transmission lines and bypass capacitors are used as bias circuits of the proposed LNA to behave as short circuits within the design bandwidth and reduce chip area. Circuit designs of the broadband drivestage and single-band stages are detailed as follows.

A. DESIGN OF THE BROADBAND DRIVE STAGE
The broadband drive stage plays a decisive role in the input matching and noise performance of the reconfigurable LNA. Inductive source degeneration is used to bring the optimum impedance for minimum NF close to the complex conjugate of input impedance. Therefore low NF and input impedance matching can be guaranteed simultaneously. Broadband input matching operation is achieved based on VOLUME 8, 2020  π-type networks. The broadband drive stage is composed of two cascaded CS 4 × 50 µm transistors. The two transistors are biased at low noise region and operate at V ds1 of 2.5 V, V gs1 of 0.6 V with drain current I ds1 of 60 mA. Fig. 9 illustrates simulation results of the broadband drive stage. The broadband drive stage exhibits a small-signal gain of 11-23 dB, a noise figure of 0.8-1.0 dB, reverse isolation of 20-35 dB with input and output reflection coefficient less than −10 dB from 8-20 GHz.

B. DESIGN OF THE TWO SINGLE-BAND LNAS
From the simulation results of the broadband drive stage, it can be found that gain variation is 12 dB from 8 to 20 GHz. In order to maintain good gain flatness in all-band mode operation, the high-band stage consists of two transistors and the low-band stage consists of one transistor. Transistors size used in high-band and low-band stages is 2 × 50 µm. Transistors are operated at V ds2 /V ds3 of 2.5 V, V gs2 /V gs3 of 0.6 V with drain current I ds2 /I ds3 of 31/15.8 mA. Fig. 10 presents simulation results of the high-band stage. The high-band stage exhibits a 15 dB small-signal gain, a noise figure of 1.0-2.0 dB, reverse isolation larger than 25 dB with input and output reflection coefficient less than −5 dB from 14-20 GHz. Simulation results of the low-band stage are shown in Fig. 11. The low-band stage presents a small-signal gain of 7-8 dB, a noise figure of 1.0-1.6 dB, reverse isolation larger than 15 dB from 8-12 GHz.
Even though every transistor is unconditionally stable, instabilities may still occur due to possible oscillation loops in the complex structure of the multi-device amplifier or due to the device's nonlinear behavior. Therefore, A R-C resistive   feedback at the gate and drain of transistor Q 4 and a resistor of 6 at the output port are used to improve the stability of the LNA. Table 1 lists a summary of total dc drain current in multi-mode operation and drain voltages states of singleband transistors that are used to realize switch-free dual-band operation. Fig. 12 presents the complete schematic of the proposed dual-band reconfigurable LNA with parameters of key devices are listed in Table 2.

IV. MEASUREMENT RESULTS
The proposed LNA is fabricated in WIN Semiconductor 0.15-µm E-mode GaAs pHEMT process. The thickness of the substrate is 100 µm. Fig. 13 shows a micrograph of the LNA with a compact 2 × 1.8 mm 2 chip size, including all RF and dc power pads. The LNA is measured by on-wafer probing at RF pads, and the dc biases of the LNA are connected to an external circuit via 25-µm diameter gold bond wires. The LNA is measured by the Cascade Microtech probe station, Summit 11000B-M. S parameters and noise performance are measured on-wafer by Agilent E5071C network analyzer and Agilent N8975B noise figure analyzer.    By turning off transistors of Q 3 and Q 4 shown in Fig. 12, the LNA operates in low-band mode from 8∼10 GHz. Fig. 15 gives the measured and simulated S-parameters,   When the transistor of Q 5 shown in Fig. 12 is turned off, the LNA operates in high-band mode from 12∼20 GHz. Fig. 16 gives the measured and simulated S-parameters and NF. LNA achieves a small-signal gain of 20.1-28 dB, NF of 1.23-1.51 dB, input and output return-loss are better than 10 dB.

B. OP1dB MEASUREMENT
Simulated and measured OP1dB of the LNA operating in different modes is shown in Fig. 17. The measured OP1dB from 8∼10 GHz is 0.5-1.7 dBm. The measured OP1dB from 12-20 GHz is 2.2-5 dBm. Table 3 compares the performance of the proposed dualband switch-free reconfigurable LNA with other state-of-theart single-band, wideband and multi-mode LNAs. The LNA presented in this article is the first switch-free reconfigurable LNA based on coupled lines. It can be found that NF of the proposed LNA is much lower than that of other reported wideband and multi-mode LNAs. Moreover, performance of the proposed LNA in different operation bands is comparable with that of the corresponding single-band LNAs. The definition of figure of merit (FOM) can be written by: