Analysis and Reduction on in-Band RCS of Fabry-Perot Antennas

In this article, a method is proposed to reduce the in-band radar cross section (RCS) of the high-gain Fabry-Perot antenna (FPA) based on the cancellation between the antenna mode RCS (AM-RCS) and structural mode RCS (SM-RCS). The ports of the two back-fed microstrip array antennas are connected to form the partial reflective surface (PRS) with a high reflectivity. The phase delay line (PDL) is proposed to control the AM-RCS to cancel out the SM-RCS. The AM-RCS is used to further reduce the in-band RCS of the low RCS high-gain FPA instead of being eliminated by the matched load. The measured results show that the FPA has a great impedance matching within the band of 9.8-11.2 GHz, and the maximum realized gain of the FPA reaches 10.7 dBi. The simulated bistatic RCS of the FPA is less than −14 dBsm within ±90 degrees angle domain at the center frequency. The minimum monostatic RCS of the proposed FPA is reduced by 17.9 dB compared with the reference FPA. Furthermore, this article provides a design basis for the feed line length of the FPA from the perspective of scattering performance.


I. INTRODUCTION
Antenna stealth technologies aim to lower the probability of antennas being detected by radar through reducing their RCS. As we all know, the RCS of antennas consists of two components: antenna mode radar cross section (AM-RCS) and structural mode RCS (SM-RCS) [1]. The AM-RCS originates from the reradiation of antennas under an incident plane wave, which is related to the radiation performance and loads of the antennas. The SM-RCS is the remainder of the total RCS to remove the AM-RCS. In the ordinary way antennas are installed on a large metal carrier, the SM-RCS much larger than the AM-RCS is the main reduction target.
The principle of RCS reduction is deflecting the scattering energy out of the threat angle or converting it into heat. Two typical methods are shaping [2] and coating radar absorbing materials [3]. The former has a limited effect and may degrade the radiation performance of the antenna, while the latter has a high cost. In recent years, antennas RCS reduction technologies based on metamaterials emerge in endlessly. The frequency selective surface (FSS) [4]- [6] and metamaterial absorber (MA) [7]- [9] can manipulate the magnitude The associate editor coordinating the review of this manuscript and approving it for publication was Shah Nawaz Burokur . of scattering waves. The former can realize the passive cancellation of scattered electromagnetic waves by arranging two frequency selective surfaces with a phase difference of 180 • into a chessboard structure, the latter can receive the incident energy into the absorber structure through spatial impedance matching, and convert it into thermal energy through the structural or resistance loss, so as to reduce the scattering energy. The phase gradient surface (PGS) influences the propagation direction of scattering waves, which is used to reduce the monostatic RCS and control the peak of bistatic RCS [10]. The polarization conversion metasurface (PCM) can scatter electromagnetic waves that are orthogonal to the polarization of incident electromagnetic waves. Besides, the PCM can be arranged as a checkerboard to realize RCS reduction [11]- [14].
The Fabry-Perot antenna (FPA) is the integrated design of antennas and metamaterials. The partial reflective surface (PRS) is used to improve the gain of the source antenna, usually a microstrip antenna. The PRS with positive reflection phase gradient can widen the gain bandwidth of the FPA [15]. By introducing the idea of phase-modulated metasurface into the design of the PRS, the beam steering of the FPA can be realized [16]- [18]. In addition, the FPA can realize partly the balance of radiation and scattering performance of VOLUME 8, 2020 This work is licensed under a Creative Commons Attribution 4.0 License. For more information, see https://creativecommons.org/licenses/by/4.0/ antennas [19]- [29]. On the one hand, the cavity composed of the PRS and reflector causes the multiple reflections of the electromagnetic waves radiated by the source antenna, which ensures the out waves in phase. On the other hand, the FPA has a good in-band scattering performance due to multiple losses in the cavity. Furthermore, metamaterials can be used to reduce the RCS outside the operating band of the FPA, thus reducing the RCS within a wide band and ensuring the stability of radiation performance. The MA replaces the metal reflector of the FPA to further improve the in-band RCS reduction, but the high gain characteristics is lost [19]. A coding metasurface is designed in the upper layer of the PRS, which realizes a broadband RCS reduction but slightly deviates the main beam direction [20]. The combination of the MA and PRS can reduce the RCS out of band and weaken the influence on the gain of the FPA [21]- [23]. The FSS and PCM are arranged as checkerboard metasurface (CM) as the upper layer of the PRS, a broadband RCS reduction can also be realized [24]- [29]. Both the PGS and MA are used to design the PRS, the PGS improve the in-band RCS and the MA reduce the RCS within a wide band [30]. In most researches, the reference antenna for RCS reduction is the microstrip antenna, which are not directly compared with FPA. In fact, the in-band RCS is more complicated than the RCS out of band, especially the AM-RCS exists in the band. The AM-RCS is not ignored under the condition that the SM-RCS can be comparable to AM-RCS for a low RCS antenna. The reflection magnitude and phase of the antenna loads have a great effect on the AM-RCS, so the radio circuit matching technology is the main method to suppress the AM-RCS of antennas. The circulator and phase shifter are terminated on the ports of an antenna array [31], [32], and the AM-RCS would be manipulated to cancel out the SM-RCS, but it requires the circulator and phase shifter with a high-level impedance matching performance. Different from the references, the AM-RCS is eliminated directly by the matched load. The RCS of a low RCS antenna is further reduced by manipulating AM-RCS in this article. The minimum RCS can be realized when the AM-RCS and SM-RCS with the equal amplitude and phase difference of 180 degrees. In this work, a method to reduce the in-band RCS of FPA is proposed based on the cancellation between the AM-RCS and SM-RCS. The transmitting and receiving microstrip antenna arrays are connected by metallized via to form the PRS of the proposed high-gain FPA. The RCS of the proposed FPA terminated with a special load is divided into the SM-RCS and AM-RCS. The feed line length of the FPA is adjusted to providing the phase difference of 180 degrees between the AM-RCS and SM-RCS at the center frequency. The impedance bandwidth of the proposed antenna is 9.8-11.2 GHz and the gain loss of the proposed antenna is 1.9 dB due to the addition of the phase delay line (PDL). The measured minimum monostatic RCS of proposed FPA terminated with open load reaches −32.7 dBsm, which is 17.9 dB lower than that of the reference FPA. At the center frequency, the bistatic RCS of the FPA terminated with open load is less than −14 dBsm within ±90 degrees angle domain, and especially it is less than −30 dBsm within ±8.7 degrees angle domain. Furthermore, this article provides the constraint condition for the feed line length of the FPA from the perspective of scattering performance.

II. THEORY OF DIVIDING SCATTERING FIELD
Assumed a linearly polarized plane wave is incident on a broadside antenna placed on the x-y plane, the schematic diagram of antenna scattering field is shown by Fig. 1. As we all know, the antennas need to be terminated with radio frequency (RF) devices such as amplifiers, filters, receivers, and so on. In fact, the RF-devices are regard as the load of the antennas, and they can not match with the antennas perfectly. Therefore, part of the energy received by the antenna is reflected by the load of the antenna, which excites the antenna again to produce a secondary radiation field in the space. The secondary radiation field is called the antenna mode scattering field (AM-SF), and the remainder of total scattering field (SF) removed to the AM-SF is the structural mode scattering field (SM-SF). Considering the relationship between the RCS and scattering field, the antenna RCS is composed of the AM-RCS and SM-RCS. The total scattering electric field E s of antennas is the vector superposition of the AM-SF and SM-SF [1].
where E s s and E s a are the SM-SF and AM-SF respectively, E t 1 is electric field radiated by the antenna excited by a unit source, b m 0 is the magnitude of the receiving electric field for the antenna with matched load, l and a are the reflection coefficient of load and antenna respectively. The E s a is related to the load and radiation field of antennas, so it matters for the high-gain antenna.
Based on the definition of AM-SF and SM-SF, the AM-SF disappears when the antenna is terminated with the matched load, and the total scattering field is equal to the SM-SF. However, the ideal matched load is difficult to achieve in practical engineering, so the method to divide the SF is proposed based on antennas terminated with two loads as following. Considering that the SM-SF independent of the loads, the SF can be divided into the SM-SF and AM-SF by calculating the total scattering electric field of antenna terminated with different loads twice, The AM-SF of the antenna with the load reflection cofficient 0 l is Without loss of generality, the open load and short load can be used to divide scattering field. The AM-SF is affected by the load, and the minimum RCS can be achieved by subtracting the AM-RCS from the SM-RCS.

III. DESIGN AND ANALYSIS OF THE ANTENNA A. DESIGN OF THE PARTIAL REFLECTIVE SURFACE
The element of the PRS is illustrated by   circular slot with radius of r 2 = 1.5 mm in the middle metal ground. Viewed from the top or bottom, the element is the y polarized microstrip antenna, so the element is transparent to the co-polarized electromagnetic waves but reflects the cross-polarized ones. The parameters p and r 2 are optimized so that the reflectivity R of the element is larger enough. The relationship between the directivity and reflectivity of PRS is where D represents the theoretical maximum directivity of the FPA excited by source antenna with 0 dBi. Fig. 3 and Fig. 4 depict the simulated reflection magnitude of the element with the change of p and r 2 . When p increases to 17 mm, the reflection magnitude reaches 0.9, whereas the period of element is larger than half wavelength of 10 GHz, which would raise the sidelobe level. The larger the parameter r 2 is, the higher the reflection magnitude is, but it would cause the resonate frequency to move down.
The final parameters are shown in Table 1.   The co-polarized reflection coefficient of the element is used to determine the height of Fabry-Perot cavity.

B. DESIGN OF THE FABRY-PEROT ANTENNA
The FPA belongs to a leaky wave antenna. Some of the electromagnetic waves radiated firstly by a source antenna are reflected in the Fabry-Perot cavity, and part of energy would be emitted through the PRS. By optimizing the cavity height, the electromagnetic waves leaked from the PRS would keep in phase to realize the high-gain FPA. In this case, the theoretical maximum directivity of the FPA excited by 0 dBi source antenna is 10.9 dBi. The cavity height is determined by the resonate condition, where ϕ up and ϕ down are the phases of the PRS and reflector respectively. λ c is the wavelength at the center frequency, h represents the height of Fabry-Perot cavity. N is generally selected as 0 to maintain the low profile of the FPA. The simulation model of the FPA is shown in Fig. 6(a). The PRS consisting of 25 elements with dimensions 75 mm ×75 mm is designed. A microstrip antenna fed by side is used as the source antenna of the FPA, as shown in Fig. 6(b). The dielectric substrate of the source antenna is  FR4_epoxy with thickness t 1 = 1 mm, and the parameters of the source antenna are: l p = 9.4 mm, w p = 6.6 mm, w 1 = 0.8 mm, w 2 = 2.1 mm, l 2 = 3.3 mm. The substrate grounded is regarded as the bottom of the Fabry-Perot cavity, and its magnitude and phase ϕ down are 1 and 154.7 degrees respectively. In this case, the height of the FPA is h = 13.9 mm, calculated by equation (5). However, there is always a deviation between theory and practice, the cavity height h needs to be optimized in order to reach the maximum broadside directivity at 10 GHz. Fig. 7 describes the simulated reflection magnitude with change of h. The PRS can slightly broaden the bandwidth of source antenna, and the operating bandwidth of the FPA increases from 5.7% to 7.4% as h increases from 13.5mm to 14.5 mm. Fig. 8 demonstrates the directivity with change of h. When h increases from 13.5 mm to 14.5 mm, the maximum directivity of the FPA increases and then decreases, and the frequency corresponding to the maximum directivity also shifts. The maximum directivity 16.4 dBi is obtained at the center frequency when h is equal to 14 mm. The maximum directivity of FPA is 9.2 dB higher than that of source of the antenna. However, the increase value of the maximum directivity doesn't reach 10.9 dB, it may contribute to the finite size of the FPA and the loss of dielectric substrate.

C. ANALYSIS OF SCATTERING PERFORMANCE
The total RCS contains the SM-RCS and AM-RCS. The former is the RCS of the antenna terminated with matched load, and the latter originates from the reradiation fields of reflected energy in the antenna port. The AM-RCS is positively correlated with antenna gain, so it needs to be more concern for high gain antenna than others. Fig. 9 demonstrates the simulated monostatic RCS of the FPA under normal incidence of plane waves with x and y polarization. The RCS curves in Fig. 9 (a) are close to coincide, so there is hardly the x polarized AM-RCS. Fig. 9 (b) shows that the y polarized RCS on short state is 4.7 dB higher than that on the open state at 10 GHz, which attributes to the AM-RCS. The bistatic RCS at 10 GHz is drawn in Fig. 10 to analyze the distribution of scattering energy. Fig. 10 (a) demonstrates that x polarized bistatic RCS is almost independent on the load. Fig. 10 (b) indicates the y polarized RCS is greatly affected by the load of the FPA, the RCS on the short state at 10 GHz is 4.7 dB higher than that on the matched state. In the end-fire direction, the RCS in the open state is larger than that in the matched state, and the difference between them is provided by the AM-RCS. Therefore, as a y polarized antenna, the AM-RCS of the FPA can be ignored under the incident wave with x polarization, while it deserves the attention for y polarization.
In order to accurately analyze the RCS of the FPA, the RCS is divided into the AM-RCS and SM-RCS based on equation (3). The RCS of the FPA is simulated by being terminated with 50 , 30000 and 0.0001 loads as the matched, open and short loads respectively. The RCS components of the FPA under normal incidence with the y polarized plane wave are depicted in Fig. 11. The magnitude and phase of the SM-RCS are always the same, since they are independent on the load states. The magnitude of the AM-RCS should be the same for the FPA terminated with loads with the same reflection magnitude, and the magnitude difference in Fig. 11 Fig. 11(c). The phase differences between the AM-RCS and SM-RCS on the open and short states are 231 degrees and 38 degrees at 10 GHz respectively. In order to reduce the RCS of the FPA, a special load should be selected to equalize the magnitudes of the AM-RCS and SM-RCS, and the microstrip feed line should be improved to realize the phase difference of 180 degrees between the two RCS components.
The load can be chosen to be close to the open load, considering that the maximum magnitude of AM-RCS is close to that of SM-RCS. The phase difference between the AM-RCS and SM-RCS is provided by the microstrip feed line except for the load. The field that a plane wave is received by the antenna and then reradiated is the AM-SF, so the excitation signal of the AM-SF needs to pass through the feed line twice. Therefore, the phase of the AM-RCS can be controlled by the microstrip phase delay line (PDL) and the VOLUME 8, 2020 The simulation model of the proposed FPA is shown in Fig. 12. The length of the PDL is l o = 11.1 mm according to equation (7) when n = 3. The final length of the PDL is optimized to 9.4 mm to minimize the total RCS of the FPA. Fig 13 describes the simulated monostatic RCS of the FPA terminated with open load under normal incidence with y polarized plane wave. Compared with the reference FPA without PDL, the monostatic RCS of the proposed FPA with PDL is reduced remarkably from −13.9 dBsm to −33.5 dBsm at 10 GHz, but the RCS of the proposed FPA is higher than that of the reference one at some frequencies in the operating band since the phase of the PDL is frequency sensitive. The monostatic RCS curves outside the operating band between the proposed and reference FPA are around coincide. The FPA can hardly receive energy outside the operating band, and the AM-RCS can be ignored.     getting the maximum gain remains unchanged, but the gain is reduced by 1.9 dB at 10 GHz. The transmission coefficient of the microstrip feed line is shown in Fig. 16. At 10 GHz, the insertion loss of the PDL is around 1.2 dB, and the phase difference between the two lines is ϕ o = 215 degrees. The gain loss of the proposed FPA is mainly caused by the PDL, and it is inevitable but can be improved by using the low lossy dielectric substrate.

IV. MEASURED RESULTS DISCUSSION
In order to verify the proposed reduction technology of in-band RCS of the high gain FPA, a 75 mm × 75 mm × 17 mm FPA is manufactured in this article, and the length of   the PDL is 9.4 mm. Four nylon columns are used to support the cavity structure. The final fabricated prototype is shown in Fig. 17. The |S 11 | of the proposed FPA is measured by the Keysight N9952A FieldFox. Fig. 18 (a) describes the |S 11 | of the proposed FPA. The measured results show that the operating bandwidth of the proposed FPA is 1.4 GHz (from 9.8 GHz to 11.2 GHz). Fig. 18  measurement setup is given in Fig. 19. The simulated and measured monostatic RCS of the FPA terminated with the open load under normal incidence of plane waves with x and y polarization are depicted in Fig. 20. When x polarized plane wave incident vertically to the proposed y polarized FPA, the monostatic RCS curve of the proposed FPA is shifted 0.1 GHz to the high frequency compared with the reference FPA.
When y polarized plane wave incident vertically to the FPA, the measured minimum monostatic RCS of the proposed FPA is −32.7 dBsm, which is 17.9 dB lower than simulated minimum monostatic RCS of the reference FPA (−14.8 dBsm). Compared with the simulated monostatic RCS curves, the trend of the measured results is consistent, but the overall curves move 0.2 GHz to the high frequency, which is caused by the inconsistency between the equivalent permittivity of substrate. After the simulation, the equivalent permittivity of FR4_epoxy used in actual processing is 4.1 instead of 4.4. Table 2 represents the comparison of the maximum in-band RCS reduction between the designed antenna and existed work. It can be observed that the Fabry-Perot cavity combined with metamaterials are used to reduce the RCS of the source antenna (microstrip antenna), but few papers have proposed RCS reduction techniques directly compared with FPA. In addition, considering that the AM-RCS is eliminated by the FPA terminated with the matched load, the metamaterials only act on the SM-RCS. In this work, we propose  the PDL to reduce the FPA without metamaterials, and the AM-RCS is used to cancel out the AM-RCS instead of being eliminated by the matched load. This work provides a new idea for RCS reduction technology of antennas, and can be combined with other RCS reduction technologies to achieve more effective RCS reduction, which will be our next work.
The AM-RCS is the important component of an antenna RCS. Compared with eliminating the AM-RCS of an antenna and treating the antenna as a general scatterer, it is more meaningful to further reduce the RCS of the antenna by utilizing the AM-RCS. The RCS reduction of the antenna can be realized by optimizing the AM-RCS, and the value of reduction is related to the magnitude difference of the SM-RCS and the AM-RCS. This article provides a new idea for RCS reduction of antennas.

V. CONCLUSION
In this article, a high reflectivity PRS is proposed to design the high-gain FPA. The microstrip phase delay line is used to control the AM-RCS phase of the FPA. The in-band RCS of the low RCS high-gain FPA is further reduced based on the cancellation between AM-RCS and SM-RCS. The bandwidth of |S 11 | < −10 dB is 9.8-11.2 GHz, and the maximum realized gain is 10.7 dBi at 10.2 GHz. The minimum monostatic RCS of proposed FPA terminated with open load is 17.9 dB lower than that of the reference FPA. The bistatic RCS of the FPA is less than −14 dBsm within ±90 degrees angle domain at center frequency. This article provides the constraint condition for the feed line length of FPA from the perspective of scattering performance, and a new idea for RCS reduction technology of antennas. He is currently the Director of the Science and Technology on Antenna and Microwave Laboratory and the Chief Expert of the Nanjing Research Institute of Electronics Technology, Nanjing. In recent years, more than 40 academic articles and five books have been published. His main research interests include phased array antenna technology, microwave circuit design, and scattering suppression technology.