Dual-Band Dual-Polarized Scalable Antenna Subarray for Compact Millimeter-Wave 5G Base Stations

A dual linearly-polarized antenna subarray that operates at 28 GHz and 38 GHz is proposed for fifth generation (5G) base stations. In contrast to earlier millimeter-wave base-station antennas that implement individual single-band antennas, simultaneous realization of dual-band operation can save space and cost. In addition, the proposed subarray depicts dual polarizations, improving signal reliability through polarization diversity. Furthermore, the proposed subarray is scalable and expandable in size and aperture. The proposed antenna subarray consists of <inline-formula> <tex-math notation="LaTeX">$2\times {2}$ </tex-math></inline-formula> dual off-center-fed dipoles. The dual-band feature is obtained by tailoring the structure for expanding current pathways and impedance bandwidths. Accordingly, the impedance bandwidths for the lower and higher bands are 27.2–30.2 GHz and 35.7–40.3 GHz, respectively. When uniformly-distributed currents are excited, the proposed antenna shows broadside radiation with peak gain of 13.1 dBi at 28 GHz and 13.2 dBi at 38 GHz. When various current phases are excited, the subarray provides a scanning range of ±18°. The scalability is demonstrated by an example large-scale array that comprises <inline-formula> <tex-math notation="LaTeX">$4\times {4}$ </tex-math></inline-formula> elements. The <inline-formula> <tex-math notation="LaTeX">$S$ </tex-math></inline-formula>-parameters are robust, and the gain is enhanced as 19.6 dBi at 28 GHz and 17.8 dBi at 38 GHz. Meanwhile, a broader scanning range of ±45° can be obtained.


I. INTRODUCTION
Recently, the fifth generation (5G) wireless communication technologies have been emerging into research fields. While both NR1 and NR2 have gathered great importance, the NR2 frequency bands, which utilize millimeter-wave wireless communication, provide more significant expansion of bandwidth and improve higher data rate. In particular, the Federal Communication Commission (FCC) has assigned both 28-GHz and 38-GHz bands [1], including n257 (26.50-29.50 GHz), n258 (24.25-27.50 GHz), n260 (37.00-40.00 GHz), and n261 (27.50-28.35 GHz), for 5G NR ecosystem development. However, millimeter-wave communication causes relatively high path loss and penetration loss [2]. To compensate these losses, the base stations must employ high-gain antennas and a digital beamforming scheme. Furthermore, the millimeter-wave 5G base stations are desired to have light weight and compact size, so the The associate editor coordinating the review of this manuscript and approving it for publication was Adnan Kavak . antenna design is required to depict high gain within a compact volume.
Earlier studies have proposed several 28/38 GHz antennas for the base stations [3]- [28]. Most of the designs aim at single bands on either 28 GHz [3]- [15] or 38 GHz [16]- [20]. Some studies propose dual-band structures that operate at both 28 GHz and 38 GHz [21]- [28]. For future millimeter-wave 5G base stations, simultaneous realization of dual-band operation can save space and cost, and thus the dual-band antenna arrays show greater potential for serving compact base stations. However, the design of dual-band arrays is challenging. The first challenge is to synthesize dual resonances on one antenna element. Even though a dualband antenna can be generated, when multiple elements form into an array, the second challenge is bandwidth reductions and frequency detuning caused by mutual coupling between elements. Furthermore, as the physical spacing between elements is fixed, the electrical spacing in terms of the 38-GHz resonance is larger. Thus, the larger electrical spacing may cause grating lobes, which reduce the antenna gain. More explicitly, the antenna performances in the literature concerning millimeter-wave 5G base stations are I summarized in Table 1. Some limitations can be observed from the earlier designs that have dual-band resonances [21]- [28]. First of all, several dual-band antenna arrays have only single polarization [21]- [23], [27], [28], but multi-polarized antenna design enables a receiver to take advantage of both the co-polarized and the cross-polarized states. Second, although some subarrays can be arranged as a large-scale antenna array, the aperture and size of the final design are fixed [21]- [28]. It is difficult to adapt these designs to base stations with arbitrary size, as a scalable or expandable feature is lacking. Third, more studies should be conducted to enhance the antenna performance in the literature. For example, the compactness of the subarray layout can be further improved [21], [22], [24], [25], and the scanning range is desired to be increased [21], [23].
The goal of this paper is to present a scalable dual-band subarray that can operate on both the 28-GHz and 38-GHz frequency bands. The distinct features of the proposed design are threefold. First of all, the proposed subarray implements dual polarizations for the dual-band operation, providing the advantage of polarization diversity that improves signal reliability. Second, the proposed subarray is scalable and expandable in size and aperture. Not only the unit element and the subarray depict desired S-parameters and radiation features, but integrated arrays with arbitrary size still provide good impedance characteristics and enhanced gain. This feature makes the proposed design provide optimum radiation performance despite the change of available antenna design space. Third, the proposed subarray provides an enhanced scanning range, which can reduce the number of required subsectors. These distinguishing characteristics can overcome the limitations in the literature.
The proposed subarray consists of 2 × 2 elements, demonstrating broad bandwidths  GHz) and high gain (13.1 dBi at 28 GHz and 13.2 dBi at 38 GHz) within a compact footprint (34.0 × 36.0 mm 2 ). The unit element comprises dual offcenter-fed dipoles [29]. By tailoring the current paths on the structure, dual-band operation with enhanced bandwidths can be obtained. Furthermore, the topology of the subarray and the separation between elements are investigated, and the optimum structure is determined in terms of S-parameters and radiation performance. To demonstrate the scalability, the proposed subarrays are integrated as a large-scale array that consists of 2 × 2 subarrays. The resultant impedance bandwidths are 26.2-28.5 GHz and 34.5-44.7 GHz, and the peak gain at 28 GHz and 38 GHz are 19.6 dBi and17.8 dBi, respectively. In addition, the envelop correlation coefficients (ECCs) and the beam scanning range are examined, and the antenna performances are compared with the state-of-the-art.

II. UNIT ELEMENT
This section describes the design, analysis, and validation of the dual-band dual-polarized unit element. Full-wave simulation is carried out using Ansoft HFSS.
A. ANTENNA DESIGN Figure 1 is the configuration of the proposed unit element, which consists of two layers separated by a 2.5-mm air spacer. The top and bottom layers are constructed on Rogers Duroid RT5880 (dielectric constant ε r = 2.2 and loss tangent tanδ = 0.0009) with a thickness of 0.5 mm and 0.8 mm, respectively. The proposed antenna is comprised of two pairs of off-center-fed dipoles above a conducting ground plane, which locates below the bottom layer and shields the antenna from the loading effect due to other electronics. The two dipoles are placed orthogonally, fed individually to excite horizontal and vertical polarization (H-pol and V-pol), respectively, where the feeding points are denoted by F 1 and F 2 . In contrast to conventional narrow-strip dipoles, which cannot provide a broadband resonance, the proposed structure applies a top hat to the ends of both arms. The top hats result in a planar dumbbell-shaped dipole [30], improving impedance bandwidths by increasing the current pathways to the top of the antenna. However, the two hats of the dipole are not realized on the same layer; instead, the two hats are constructed on separated layers. The feed line can be implemented through a 50-Sub-Miniature Pushon Micro (SMPM) connector [31], which depicts advantages of compact size, reducing system weight while offering high reliability.
The antenna geometry is designed and optimized to generate the dual-band resonances. For the H-pol (V-pol) dipole, the geometric parameters include the length and width of the top-loaded hat, L H 1 (L V 1 ) and W H 1 (W V 1 ), respectively, and the length and width of the central narrow strip, L H 2 (L V 2 ) and W H 2 (W V 2 ), respectively. As an offset feeding scheme is adopted, the H-pol and V-pol dipoles are asymmetric. To further improve the dual-band impedance bandwidths, a pair of stair-wise patch is integrated to one of the hat of the V-pol dipole, serving the antenna as an additional impedance matching technique. The finalized geometric parameters are  To explain the mechanism of two resonances, the surface current distributions on the proposed antenna are presented in Figure 3. The overall current distributions can be evaluated as the sum of the travelling-wave currents and the standing-wave currents [32]. Considering the H-pol dipole at the resonance of 28 GHz, at t = 0 and t = T /4 (T denotes the period of oscillation at each frequency), the maximum current amplitude occurs at the offset feeding point, F 1 , instead of the phase center of the dipole, which is the current maximum of a conventional center-fed dipole. On the other hand, the minimum current amplitude is at the opposite location to the feeding point, A. This indicates that the traveling-wave currents are more significant while the standing-wave currents are weak. For the resonance at 38 GHz, the currents depict the maximum at the feeding point, F, for both times. At t = 0, the currents show the minimum amplitude around A; however, at t = T /4, the currents with the maximum amplitude flow around the edge of A. This indicates that the traveling-wave and standing-wave currents are revealed alternatively. As this unit element provides two current pathways with different lengths, combining the two resonances enable multiband operation. Thus, the design rule of this antenna is to adjust the geometric parameters of the structure, namely, L H 1 , L H 2 , and W H 1 , to control the current flowing through the upper and lower sides of the dipole.

B. PERFORMANCE
The proposed antenna structure is fabricated and tested. Figure 4 is the photograph of the test pieces for the dual-band antenna element. The impedance matching is shown in Figure 5. The simulated impedance bandwidths are 27.6-30.8 GHz and 35.4-38.9 GHz, and the measured ones are 27.8-29.9 GHz and 36.1-38.3 GHz. Dual-band impedance matching is achieved, and the simulated results agree well with the measured ones. Moreover, the isolation between the H-pol and V-pol feeding ports is presented in Figure 6. At the resonance of 28 GHz, the simulated and measured isolation are 15.4 dB and 14.3 dB, respectively. At 38 GHz, the simulated and measured isolation are 22.0 dB    and 17.4 dB, respectively. Although the isolation between the H-pol and V-pol ports is about 15 dB at 28 GHz, this isolation can be enhanced using signal processing algorithms at a baseband or intermediate-frequency (IF) level [33]- [36]. As multipath propagation, rainfall, equipment imperfections, and antenna misalignment degrade the isolation between the H-pol and V-pol signals, state-of-the-art MIMO communication systems must implement interference cancellation techniques for the H-pol and V-pol mutual coupling. For example, Kalman-based algorithms can jointly recover the phase of both the co-polarized signal and the interfering one using carrier phase synchronization [33], [34]. An algorithm based on soft constraint satisfaction (SCS) is proposed to eliminate mutual coupling between the H-pol and V-pol signals using a digital signal filter at the baseband [35]. A joint equalizer is proposed to deal with the mutual coupling between H-pol and V-pol ports and inter-symbol interference (ISI) simultaneously based on Least Mean Square (LMS) [36]. These techniques can create more significant improvements of isolation as compared to Radiofrequency (RF) implementation alone.
The radiation characteristics of the unit element are also measured. Figure 7 exhibits the realized gain patterns. At the resonance of 28 GHz, the simulated and measured broadside gain are 6.9 dBi and 5.7 dBi, respectively. On the x-z plane, the simulated and measured broadside cross-polarization discriminations (XPDs) are 25.1 dB and 24.2 dB, respectively. On the y-z plane, the simulated and measured results become 23.4 dB and 24.1 dB, respectively. At 38 GHz, the simulated and measured broadside gain are 5.3 dBi and 5.5 dBi, respectively. The simulated and measured broadside XPDs on the x-z plane are 22.9 dB and 20.0 dB, respectively. On the y-z plane, the simulated and measured results become 21.6 dB and 22.8 dB, respectively. Thanks to the back conductor, the unit antenna element depicts broadside radiation at both the resonant frequencies. Figure 8 provides the simulated and measured antenna efficiency, which include radiation efficiency as well as impedance matching. The simulated (measured) efficiency at 28 GHz and 38 GHz are 86.5% (88.8%) and 85.4% (69.7%), respectively. Within the operating impedance bandwidth, the measured efficiency is generally larger than 50%, which is sufficient for a millimeter-wave application. Finally, the simulated and measured realized peak gain are provided in Figure 9. The simulated and measured realized peak gain at 28 GHz are 7.4 dBi and 7.8 dBi, respectively, whereas the simulated and measured results at 38 GHz are 7.7 dBi and 5.5 dBi, respectively. These impedance and radiation characteristics validate the proposed dual-band dual-polarized antenna element.

III. ANTENNA SUBARRAY A. ANTENNA DESIGN
This section describes a scalable subarray that consists of 2 × 2 identical unit elements. Figure 10 presents the  The proposed quasi-triangular lattice can delay the existence of grating lobes and miniaturize the overall dimensions.
The proposed subarray implements 8 ports, where ports 1, 3, 5, and 7 excite H-pol radiation and ports 2, 4, 6, and 8 excite V-pol plane wave. The currents that feed these ports can be controlled by a digital beamforming module. Although the dimensions of the circuit board are L 2 = 80 mm and W 2 = 80 mm, the subarray takes up space with dimensions of only 34 × 36 mm 2 .
To investigate the operating principle, Figure 11 presents the simulated surface current distributions of the proposed antenna subarray, where the H-pol ports are excited in phase. The current distributions of the element in the subarray are similar to the case of the unit element. For the resonance at 28 GHz, the maximum and minimum of current amplitude are the offset feeding point and the opposite location of the feed, respectively, which correspond to the resonance principle of the off-center-fed dipole. For the resonance at 38 GHz, the traveling-wave and standing-wave currents are appeared alternatively. In addition, the four elements depict similar current distributions. Accordingly, when the four elements are excited in phase, these current distributions are expected to create broadside radiation at both the frequencies. Major challenges to dual-band scalable subarrays are the avoidance of grating lobes and robust S-parameters. To demonstrate that the proposed subarray geometry can elude grating lobes, two reference configurations in a standard triangular lattice are analyzed. The geometry of the two reference designs is shown in Figure 12. Conventionally, an array in a rectangular lattice leads to a larger separation VOLUME 8, 2020 between elements; in contrast, the arrangement in a triangular lattice can make use of the design space more efficiently [37]. For the two reference subarrays, denoted by Ref1 and Ref2, the separations between the elements are d x = d y = 10.0 mm and d x = d y = 9.0 mm, respectively, where d x and d y are the distance between the phase centers of adjacent elements. Ref2 employs a closer separation, which leads to a more compact design and was expected to elude the occurrence of grating lobes.
First of all, the impedance matching of Ref1, Ref2, and the proposed subarray are presented in Figure 13. These results are evaluated by exciting port 1, while the other ports are terminated to match load. Both the dual bands can achieve robust matching using the three array arrangements. However, as the radiation patterns at 38 GHz are tested, Ref1 and Ref2 exhibit significant grating lobes. The four H-pol ports of the three subarrays are excited, and the resultant y-z plane patterns at 38 GHz are shown in Figure 14.
Although the three subarrays provide a local maximum at the broadside direction, grating lobes occur at about ±50 • . Concerning Ref1, peak gain of 10.7 dBi appears at -63 • , whereas the broadside gain is 4.5 dBi. Concerning Ref2, peak gain of 8.3 dBi appears at −41 • , whereas the broadside gain is 5.9 dBi. These results indicate that the reference subarray geometry fails to prevent the occurrence of grating lobes. In contrast, the proposed subarray preserves the main beam to be directed broadside. The first side lobe level is 2.1 dBi, at −73 • , which is lower than the broadside gain of 9.5 dBi. As a result, the proposed subarray in the quasi-triangular lattice overcomes the challenge of dual-band array design.

B. PERFORMANCE
The S-parameters of the subarray are tested and compared with the results of the unit element. The photograph of the test piece is presented in Figure 15. The simulated and measured impedance matching of port 1 are shown in Figure 16 The measured impedance bandwidths are slightly narrower than the simulated ones, but the dual-band feature is achieved. In addition, the simulated impedance matching at ports 1, 3, 5, and 7 are exhibited in Figure 16(b). The four results are highly comparable, demonstrating sufficient impedance matching at these ports. Please note that the simulated result of port 1 in Figure 16(a) is slightly different from those in Figure 16(b). In Figure 16(a), ports 2-7 are terminated to open load; on the other hand, in Figure 16(b), the other ports are terminated to match load. The simulated and measured isolation are shown in Figure 17(a). For the isolation between ports 1 and 2, the    Additionally, the comparison of isolation for various ports is presented in Figure 17(b). The lowest isolation is the one between ports 1 and 2. Otherwise, the in-band isolation is generally greater than 20 dB. Figure 18(a) presents the simulated and measured realized gain patterns when port 1 is excited and the other ports are terminated to open load. These results examine the agreement between simulated and measured far-field radiation characteristics. At 28 GHz, the simulated and measured broadside XPDs on the x-z plane are 21.1 dB and 18.3 dB, respectively. On the y-z plane, the simulated and measured broadside XPDs are 17.4 dB and 18.1 dB, respectively. At 38 GHz, the simulated and measured broadside XPDs on the x-z plane are 14.4 dB and 26.9 dB, respectively. On the y-z plane, the simulated and measured broadside XPDs are 16.0 dB and 16.1 dB, respectively. The trends of the two approaches are comparable. Figure 18(b) shows the simulated gain patterns when all the H-pol or V-pol ports are excited by uniformly-distributed currents. At the resonance of 28 GHz, the H-pol and V-pol peak gain are 13.2 dBi and 9.5 dBi, respectively. At 38 GHz, the H-pol and V-pol peak gain are 10.5 dBi and 13.2 dBi, respectively. These results suggest a high-gain feature. It is noted that the proposed subarray is not tested with extra feeding networks, for digital beamforming will be employed to excite the array in future 5G base stations; thus, the measured radiation performance neglect the loss of the feeding network.
Another important performance index for 5G base stations is the envelop correlation coefficient (ECC) between ports i and j, denoted by ρ ij . The ECC can be expressed in terms of the S-parameters [38]: This expression provides predictions in several experimental environments, especially indoor environments with rich multipath propagation, which are the aim of the 5G base stations. The ECCs of the proposed antenna subarray calculated by the simulated S-parameters are shown in Figure 19(a). To confirm the validity of the results, the ECCs are also evaluated using the radiation patterns: where · denotes a Hermitian inner product and F i (θ, ϕ) is the field pattern of the antenna array when only the port i is excited and all other ports are terminated to match load. Figure 19(b) presents the calculated results, which agree well with the ECCs computed using the S-parameters. The ECCs of the proposed subarray are less than 0.03 for the dual bands, meeting the requirement of base stations that use multiple antennas. Finally, the scanning range of this subarray is tested. The four elements are fed by currents with progressive phases, which can shift the direction of main beam. The resultant gain, pattern, and pointing angle of the main beam are recorded. In general, the highest gain occurs when the pointing angle is broadside. The wider the pointing angle, the smaller the gain is; accordingly, the difference between the gain and the maximum achievable gain is defined as scan loss. The scanning range is further defined as the range of angles that lead to 3-dB scan loss. By testing potential  combinations of the progressive phases, the beam scanning performance for the subarray is depicted in Figure 20. The scanning ranges at 28 GHz and 38 GHz are ±18 • . Such beam scanning performance and the isolation between ports are influenced by the separation between unit elements. Our design strategy is to select a separation that prevents grating lobes. The isolation can be enhanced using baseband and IF interference cancellation techniques, whereas the HPBW and the beam scanning range can be improved using a larger-scale antenna array.

IV. SCALABILITY
As compared to the earlier studies regarding millimeter-wave 5G base-station antennas [21]- [28], a distinct feature of the proposed subarray is scalability in aperture and size. This section demonstrates the scalability by an example that consists of 2 × 2 identical subarrays. Figure 21 is the geometry of the large-scale antenna array. This antenna array consists of 16 unit elements, which result in 32 ports. The separations between unit elements are identical as those described in Figure 11. Ports with odd numbers excite H-pol radiation, whereas even numbers ports excite V-pol plane wave. The dimensions of the large-scale antenna array are 66.5 × 70.0 mm 2 . The robustness of S-parameters and radiation features are tested. Figure 22 presents the comparison of simulated reflection coefficients for the unit element, the port 1 of the subarray, and the various ports of the large-scale array. The results are almost identical, and both the dual bands indicate good and robust impedance matching. Figure 23 shows the realized gain patterns when all the 16 H-pol elements are excited with uniformly-distributed currents. The main beam of the antenna array is directed broadside. The peak gain at 28 GHz and 38 GHz are 19.6 dBi and 17.8 dBi, respectively. In particular, at 28 GHz, the halfpower beam widths (HPBWs) on the x-z and y-z planes are 12 • and 11 • , respectively. At 38 GHz, the HPBWs on the x-z and y-z planes are 10 • and 11 • , respectively. These results indicate that the radiation performance can be enhanced by straightforwardly integrating the proposed  subarrays. As the FCC regulation allows 28-GHz base stations to transmit at maximum effective isotropic radiated power (EIRP) of 73 dBm per polarization [39], [40], the maximum power handling of the large-scale antenna array is 53 dBm. When the proposed subarray is expanded into even larger scales, this maximum power is varied over the number of elements as well as the overall antenna gain, subject to the peak EIRP regulation. In addition, the sidelobe level (SLL) at 38 GHz is about -14 dB, which is a typical value for uniformly-excited current distributions. This SLL can be improved using two schemes. First, the SLL inversely varies as the number of elements, so an even larger-scale array will not depict a SLL as high as that for this array. Second, in general, the excitation scheme of a beamforming integrated circuit (IC) does not implement uniform distributions; in contrast, Dolph-Tchebyscheff excitations can improve the SLL based on an acceptable threshold. VOLUME 8, 2020 Figure 24 presents the scanning range of the large-scale antenna array. The progressive phase of the feeding current is controlled so that the main beam can be pointed at various angles. The scanning ranges at 28 GHz and 38 GHz are about ±45 • . By arranging the array on the surface of an octagonal prism structure, the azimuth plane can be served without communication deadzones. These characteristics validate the scalability of the proposed subarray. Although the design space of 5G base-station antennas varies over different models, the proposed subarray can exploit an available area and provide enhanced radiation performance. After validating the performance of the large-scale antenna array, the proposed antenna can serve MIMO antenna systems as a 5G beam-forming unit. A scenario of the deployment of the proposed antenna is shown in Figure 25. An input signal is assigned and transmitted to a 16-way power divider. The output of the power divider is cascaded with a beamforming IC which consists of four H-pol and four V-pol channel radiofrequency (RF) control. Each channel comprises RF components including circulator, variable attenuator, variable phase shifter, high power amplifier (HPA), driver, limiter, and low noise amplifier (LNA), providing the ability to steer a radiation pattern. The four H-pol and V-pol channels are connected to the odd numbers and even numbers of ports of the proposed subarray, respectively. The kernel that controls the feed conditions for the antenna is the variable attenuator, which provides 32 dB attenuation range with 0.5 dB per step, and the variable phase shifter, which offers 360 • phase shift coverage with 5.625 • per step. By assigning the bit combination for the attenuator and the phase shifter, beamforming can be performed based on the signal processing results. The output port of the beamforming IC is terminated with a K-type connector. For this scenario that employs 16 beamforming ICs, the 16 RF signals are combined and controlled through the 16-way power divider. Such power dividers can be implemented using multiple stages of 2-way Wilkinson power divider structures [41], [42], which use two output ports extended and coupled through a coupled line section. Furthermore, as the proposed antenna subarray is scalable, the overall array size can be expanded. Additional antenna ports are readily connected to the channels of other beamforming ICs, so the proposed scheme can serve the MIMO antenna system that consists of 64 H-pol and 64 V-pol channels.

V. CONCLUSION
In this paper, a dual-band dual-polarized antenna subarray with scalability in aperture and size for 5G compact base stations is presented. The unit element is designed as the dual off-center-fed dipoles, which operate in 27.6-30.8 GHz and 35.4-38.9 GHz with broadside gain of 6.9 dBi at 28 GHz and 5.3 dBi at 38 GHz. By combining 2 × 2 unit elements as the subarray, the proposed antenna exhibits three distinct features. First of all, the proposed subarray provides 4 horizontal and 4 vertical feeding ports with low ECC (ρ ij < 0.03). This mechanism can enhance transmission and reception efficiency using polarization diversity. Second, although the design space is limited to 34.0 × 36.0 mm 2 , the proposed subarray provides high gain (13.1 dBi at 28 GHz and 13.2 dBi at 38 GHz), broad bandwidths (27.2-30.2 GHz and 35.7-40.3 GHz), broadside radiation, and low ECC under uniformly-distributed currents. Third, the proposed subarray can be expanded into large-scale arrays with arbitrary size. This paper demonstrates the scalability through a 4 × 4 antenna array that comprises identical unit elements. Robust S-parameters and enhanced radiation features are identified. The peak gain becomes 19.6 dBi at 28 GHz and 17.8 dBi at 38 GHz, and the scanning ranges of the dual bands are ±45 • . These results validate the proposed subarray for compact millimeter-wave 5G base stations.