A Simple High-Gain Millimeter-Wave Leaky-Wave Slot Antenna Based on a Bent Corrugated SIW

We propose a bent corrugated substrate integrated waveguide (BCSIW) structure that can be used to design a high gain leaky wave antenna (LWA). The design removes the need for a metallic via fabrication process needed for standard substrate integrated waveguides (SIW) and displays superior performance to previously reported structures. We use simulations to compare the performance of the proposed BCSIW LWA to equivalent standard SIW and corrugated SIW (CSIW) structures, as well as experimentally characterize a fabricated BCSIW LWA. Simulation results show that the BCSIW structure can help improve the impedance bandwidth of a slotted LWA by about 14.7% while still maintaining high gain (about 13.2–17.4 dBi) as compared to an LWA based on a CSIW structure. Measurement results indicate that the proposed BCSIW LWA has a wide impedance bandwidth (32.6%) and a high peak gain (12–16.2 dBi) throughout a large frequency range from 22 to 29.2 GHz with a large beam angle range from −69° to −10°.


I. INTRODUCTION
In 2016, the Federal Communications Commission (FCC) announced the operating frequency spectrum for the development of wireless technology towards fifth generation (5G) technology, which included the millimeter-wave (mmW) operating band 27.5-28.35 GHz for 5G communication [1]. The International Telecommunication Union (ITU) has similarly identified the band from 24. 25-27.5 GHz for 5G communication [2]. However, currently available millimeter-wave devices are in general not commercially viable because of severe attenuation and limitations due to e.g. line-of-sight issues. Thus, there is a need for antennas that have a low profile, low cost, a compact structure and high gain to enable effective 5G communication devices and wireless networks [3]. This has in recent years The associate editor coordinating the review of this manuscript and approving it for publication was Weiren Zhu . promoted a large amount of work aimed at rapid development of planar leaky-wave antennas (LWAs) based on various types of radiation structures [4]- [6]. Among these planar LWAs, substrate integrated waveguide (SIW) has attracted much interest due to advantages such as low loss, easy fabrication and cost-effectiveness [7]- [10].
SIW-based planar one-dimensional (1D) LWAs can be classified into two groups: uniform (or quasi-uniform) and periodic. Uniform LWAs have a periodic radiator structure with a period much smaller than the guided wavelength and in which the guided wave is a fast-wave with respect to the free space. The phase constant is smaller than in free space, meaning that these fast-wave LWAs are able to radiate directly. An example from the literature of uniform LWA devices is a half-mode substrate integrated waveguide leaky-wave antenna (HWSIW-LWA) with wide impedance bandwidth and a quasi-omnidirectional radiation pattern [11]. By etching straight long slots, periodic transverse slots and VOLUME 8, 2020 This work is licensed under a Creative Commons Attribution 4.0 License. For more information, see https://creativecommons.org/licenses/by/4.0/ periodic H-shaped slots on the top of uniform SIW structures it is possible to achieve low sidelobe levels, wide impedance bandwidth as well as narrow beams and circular polarization [12]- [14]. Periodic LWAs have a radiator structure with a period that is comparable to the wavelength of the guided wave and thus periodic LWAs typically support a non-radiating fundamental mode (a slow-wave with respect to the free space) while the first-harmonic mode is in the fast-wave region, i.e. periodic LWAs radiate from the first-harmonic mode. Periodic −45 • slots, periodic antipodal tapered slots and T-shaped transverse and longitudinal slots etched on the top of periodic SIW structures have been shown to being able to produce effects such as linear and circular polarization as well as broadside scanning performance [15]- [17].
To replace metallic vias while maintaining the advantages of the SIW, Chen et al. proposed a corrugated SIW (CSIW) structure, which uses open-circuit quarter wavelength microstrip stubs in lieu of metallic vias in order to artificially create electric sidewalls [18]. Such a structure supports the TE 10 mode in the same manner as an SIW structure does [19]. In comparison with an SIW structure, the CSIW maintains the DC isolation between the top and bottom conductors by use of the open-circuit quarter wavelength stubs instead of metallic vias. CSIW structure enables tunable leaky-wave antennas, and a beam scanning antenna using an electronically controlled LWA based on a CSIW was recently proposed [20]. In order to reduce the length of the quarter-wavelength open-stubs, a half-mode CSIW structure utilizing bent (or fan-shaped) quarter-wavelength open-stubs was recently proposed [21], however, the design suffers from drawbacks such as small scanning angle range, complex structure and fabrication, as well as low gain.
In this paper we introduce an LWA based on a whole-mode bent corrugated SIW (BCSIW) with periodic transverse slots. We analytically compare the performance of our proposed structure with equivalent SIW and CSIW LWAs, and we experimentally analyze a fabricated BCSIW LWA. The use of bent open-circuit quarter wavelength stubs results in a wider impedance bandwidth while maintaining high gain and low sidelobe levels as compared to SIW and CSIW LWAs.

II. DESIGN AND COMPARISON OF SIW, CSIW AND BCSIW LWAs
As discussed in the introduction, unlike a conventional SIW structure, a CSIW structure creates electrical sidewalls using open-circuit quarter-wavelength microstrip stubs in lieu of metallic vias in the SIW structure. Utilizing bent microstrip stubs will further increase the performance as as will be shown in the following. Equivalent SIW, CSIW and BCSIW structures are shown in Fig. 1(a)-(c), respectively, and corresponding simulated electrical field distributions at 25 GHz are shown in Fig. 1(d)-(f), where it can be clearly seen that all structures support the same TE 10 mode. It is thus feasible to design periodic BCSIW LWAs utilizing the same theoretical basis as for periodic SIW LWAs. As indicated in Fig. 1 (c), the bent open-circuit quarter wavelength stubs can be divided into three parts with lengths L 1 , L 2 and L 3 . The total length, L 1 + L 2 + L 3 , should be about one quarter of the guided wavelength. The specific lengths of L 1 , L 2 and L 3 through simulation by minimizing the return loss and insertion loss of the BCSIW. As an example of a simple and effective radiator, rectangular slots are etched on the upper surface of the waveguide [22] to enable the LWA radiate. The required slot length can be estimated by where λ 0 is the wavelength at the operating center frequency in free space, and ε r is relative permittivity of the antenna substrate [23]. We choose, in accordance with [23], the slot width, w, to satisfy w/l 1. For a uniform LWA, the phase constant, β, is smaller than the wavenumber in the free space k 0 , and the radiation angle θ is defined by However, for a periodic LWA, β is larger than k 0 and thus a periodic LWA should be designed to allow the first space harmonic (n = −1) to radiate instead of the fundamental harmonic, i.e. the radiation angle is defined by where β −1 is the phase constant of the n = −1 space harmonic. The phase constant of the nth space harmonic is defined as where p is the period of the structure. In the first space harmonic mode, the periodic LWA can radiate in either the forward direction or the backward direction.
In the below we optimize the design of SIW, CSIW and BCSIW LWAs through numerical simulations. ANSYS HFSS 18.0 simulation software is used throughout.
The overall geometry of the periodic BCSIW LWA is depicted in Fig. 3(a). Taconic TLY-5-0200 for the substrate; it has a thickness of h = 0.51 mm, a dielectric constant of ε r = 2.2 and a loss tangent of tanσ = 0.0009. L × W are the total dimensions of the BCSIW LWA and P slot is the period of the slots. Plastic screws with diameter D v , placed a distance P v apart, are used to fix the LWA. Fig. 3(b) details a microstrip taper, which is designed for the antenna feeding line. The use of a tapered microstrip enables the BCSIW waveguide to a 2.92 mm end-launcher with better impedance matching for the antenna input port as compared to a traditional strip line. D e is the diameter of the assembly holes for the 2.92 mm end-launcher and P e is the distance between the two assembly holes at either side of the end-launcher. L 50 × W 50 are the dimensions of the 50 microstrip feeding line. L t is the length of the tapered structure and W t is the width of the wide side of the tapered structure. Fig. 3(c) shows detailed notations for a unit cell of the BCSIW LWA. The dimensions of the slot are L slot × W slot . The width of the waveguides is a. All the stubs have the same width W stub , and p is the gap distance between the short part of one bent stub and the long part of an adjacent bent stub.
We compare the performance of the BCSIW LWA with equivalent designs of SIW and CSIW LWAs, the unit cells of which are depicted in Fig. 3(d) and (e). The tapered microstrip structure for the coupling to the SIW and CSIW LWAs is the same as for the BCSIW LWA. The diameter of the metallic vias d and the period of metallic vias p are designed according to SIW theory [8]. The length and width of the CSIW LWA microstrip stubs are L stub × W stub . For the SIW and CSIW LWAs, the dimensions and the period of the slots are kept the same as for the BCSIW LWA for objective comparison of the radiation characteristics of the three structures. We note that  the period p of the SIW metallic vias and the period of CSIW microstrip stubs are the same. We summarize the parameters of the SIW, CSIW and BCSIW LWAs in Table 1.
Simulated S-parameters of the SIW, CSIW and BCSIW LWAs are shown in Fig. 4, where it can be seen that S 11 is below −10 dB from 21.45 to 29.65 GHz for the BCSIW LWA while from 23.8 to 28.85 GHz for the SIW LWA and from 21.95 to 29.1 GHz for the CSIW LWA; i.e. the −10 dB impedance bandwidth of the BCSIW LWA is 62.3% larger than that of the SIW LWA and is 14.7% larger than that of the CSIW LWA. S 21 is lower than −4.0 dB in the impedance band (defined as the frequency range where S 11 is below −10 dB) for the BCSIW LWA while lower than −4.4 dB for the SIW LWA and lower than −3.3 dB for the CSIW LWA. Fig. 5 shows the simulated realized gain of the SIW, CSIW and BCSIW LWAs at various frequencies in the BCSIW LWA impedance band. We note that the realized gain of the SIW LWA is −3.33 dBi at 22 GHz, i.e. the SIW LWA cannot radiate effectively at 22 GHz. We furthermore notice that the realized gain of the BCSIW LWA on average is higher than that of the SIW LWA and CSIW LWA in the BCSIW LWA impedance band. Fig. 6 shows simulated normalized radiation patterns of the co-polarized fields in the yz-plane of the SIW, CSIW and BCSIW LWAs at 22, 23.2, 24.4, 25.6, 26.8, 28 and 29.2 GHz. The sidelobe level of the three LWAs are overall similar, however it is better for the BCSIW LWA than for the SIW and CSIW LWAs at 22 GHz. Though all three LWAs have the same size fixed radiating elements, the radiation beam angle of the SIW LWA is somewhat tilted compared with those of the CSIW and BCSIW LWA, which is due to the microstrip stubs. Due to the bent microstrip stubs, the impedance bandwidth of BCSIW LWA is larger than those for the other two LWAs, and consequently the total scan angle within the impedance bandwidth of the BCSIW LWA is larger than for the other two LWAs. Within the respective impedance bandwidth of the three antennas, the total scan angles are found to be:  Table 2 compares the impedance bandwidths of the three LWAs. Each frequency point where S 11 is below −10 dB, is ticked. The scan angles corresponding to the −10 dB impedance bandwidths are shown in the last row and we can see that the BCSIW LWA has both the largest bandwidth and the largest scan angle.

III. EXPERIMENTAL CHARACTERIZATION OF THE BCSIW LWA AND COMPARISON WITH SIMULATIONS
The proposed BCSIW LWA was fabricated (as shown in Fig. 7) and measured for verification. The two ports of the BCSIW LWA are connected by two end launch connectors,

TABLE 2.
Impedance bandwidth performance summary of the three LWAs. Frequency points where S 11 is below −10 dB are ticked. The corresponding scan angles are shown in the last row. and S 11 was measured using an Agilent N5247A network analyzer. Measured S-parameters are shown in Fig. 8 together with the previously simulated values, both of which is less than −10 dB in the band of interest (i.e. from 22 to 29.2 GHz). The measured impedance bandwidth 21.3 -29.6 GHz is slightly wider than the simulated result (21.45 -29.65 GHz), which we attribute due to fabrication error. Both the measured and simulated results show that S 11 is less than −10 dB   in the band of interest (i.e. from 22 to 29.2 GHz). The measured S 21 is in the range −7.1 to −22.0 dB, which is lower than the simulated S 21 (−4.0 to −15.6 dB). The discrepancy between the simulated and measured S 21 , can, in parts, as well be attributed to fabrication error, however also due to loss in two adaptors, which have an insertion loss of 0.05 × √ f (GHz) (dB), that are used to connect the BCSIW LWA to the two ports of the network analyzer. Fig. 9 shows the setup of the microwave anechoic chamber test, in which one port of the antenna is terminated by a 50 load for the gain measurement. Fig. 10 shows the simulated and measured realized gains of the BCSIW LWA at different frequencies. The measured realized gain agrees well with the simulated result with a variation of less than 2.3 dBi, which is attributed to losses in the coaxial waveguide and  fabrication error. In the range 22 -29.2 GHz, the maximum realized gain is 16.2 dBi at 29.2 GHz. Simulated and measured normalized radiation patterns of the co-polarized fields are illustrated in Fig. 11. The measured normalized radiation patterns show that the scan angle of the fabricated BCSIW LWA can be swept from −69 • to −10 • when the frequency is scanned from 22 to 29.2 GHz, which again agree well with the simulated results. Furthermore, the sidelobe levels of the measured radiation patterns at chosen frequencies are all below −10 dB. Although there is a shift between the measured and simulated sidelobe level due to cable losses and fabrication errors, the measured and simulated curves overall agree well with each other. Use of better adaptors and connectors with lower loss, and higher precision fabrication are expected to improve the results.

IV. COMPARISON WITH PREVIOUS WORK
In Table 3 we compare the pertinent parameters and performance between our proposed BCSIW LWA and previously published LWAs and we can see that the bandwidth, scanning range and the measured gain of the BCSIW LWA are better than those in Refs [13], [24], [25] and [27]. Comparing to the SIGW LWA in Ref [26], the measured bandwidth, scanning range and length (compactness) of the BCSIW LWA are much better, though the measured gain is slightly smaller.

V. CONCLUSION
In this paper, a BCSIW LWA for millimeter-wave applications has been proposed and compared with equivalent SIW and CSIW LWAs using simulations. Here the specific shape is just as a simple and natural example to illustrate the principle how bent stubs in CSIW can improve the performance of an LWA.The BCSIW LWA is also fabricated and characterized. The measured impedance bandwidth reaches 32.6% around the center frequency of 25.45 GHz with a measured peak gain of 16.2 dBi at 29.2 GHz and a smaller than −10 dB sidelobe level. Furthermore, the scan angle ranges from −69 • to −10 • when the frequency is scanned from 22 to 29.2 GHz.
By bending the open-circuit quarter wavelength stubs of an equivalent whole-mode CSIW LWA, the total width of the BCSIW LWA is reduced by about 10.4%. Compared to the equivalent CSIW LWA, the BCSIW LWA impedance bandwidth (S 11 below −10 dB) is about 14.7% larger. Good agreement between the simulated and measured results of the BCSIW LWA has been demonstrated. Having the advantages of easy fabrication, wide impedance bandwidth and high gain, the BCSIW LWA proposed in this paper has strong potential for use in 5G communication applications. SAILING HE (Fellow, IEEE) received the Licentiate and Ph.D. degrees in electromagnetic theory from the Royal Institute of Technology, Stockholm, Sweden, in 1991 and 1992, respectively. He has been with the Department of Electromagnetic Engineering, Royal Institute of Technology, as an Assistant Professor, an Associate Professor, and a Full Professor. He is currently a Professor with the Centre for Optical and Electromagnetic Research, Zhejiang University, China. He has first-authored one monograph (Oxford University Press) and has authored or coauthored more than 600 articles in refereed international journals. He has given many invited/plenary talks in international conferences, and has served in the leadership for many international conferences. His current research interests include electromagnetic metamaterials, optoelectronics, sensing, and communication.