Low Loss W-band Packaged Filtering Balun Based on a Modified Quarter-mode Folded Substrate-integrated Waveguide Cavity

A novel packaged filtering balun is proposed by modifying quarter-mode edge and diagonally folded SIW (FSIW) cavities. The TM220 mode is selected to realize the differential signal conversion by using a quarter-mode diagonally FSIW. Furthermore, TM130, and TM330 modes are obtained by using a quarter-mode edge FSIW cavity by analyzing the mode field distributions. A W band packaged filtering balun is designed and simulated. Then, it is fabricated with 70 um thickness per layer based on our in-house Si-based MEMS photosensitive film process. The size of the filtering balun is only 2.83 λg2 with insertion loss of 3 + 1.8 dB. These types of packaged filtering balun are very suitable for future 3D heterogeneous integration with other semiconductor devices.


I. INTRODUCTION
A component that realizes the conversion between unbalanced and balanced ports, plays an important role in a millimeter wave communication system. Filtering balun not only provides the function of suppressing common mode signals to obtain differential feeds but also offers signal transmission in the passband and attenuation characteristics in the stopband. Often, the filtering balun needs to be compatible with the packaging technology, which is characterized by easy integration, planarization, and small volume.
Traditional filtering baluns are generally based on Marchand balun [1]- [2]. They may have some advantages in bandwidth, but their insertion loss is usually large [3]- [5], especially in the millimeter-wave band. Substrate integrated waveguide has its advantages of low loss at millimeter wave, high power handling capability, and compatibility with planar circuits. The design of the filtering balun in a substrate integrated waveguide relied on the application of the cavity mode which produced phase difference of 180° [ 6]- [8]. However, to the best of our knowledge, most of the filtering balun based on mircostrip techniques were operated in microwave band. Few study have demonstrated in a quarter-mode folded SIW (FSIW) cavity in millimeter wave. Quarter-mode FSIW offers the advantages of less spurious modes and at least 75% size reduction compared with standard SIW cavity.
In this paper, a multi-layer packaged filtering balun is proposed by modifying a quarter-mode FSIW cavity, thereby making its size more compact than those of many other designs. The modes where quarter-mode edge and diagonally FSIW cavities can be precisely controlled are investigated. The W-band multilayer filtering balun was fabricated by adopting benzocyclobutene (BCB) as dielectrics. BCB is a photosensitive polymer that offers the advantages of a low dissipation factor and a low dielectric constant. The fabrication method for larger BCB thickness has been investigated to improve the quality factor of the cavity, which will result in lower insertion loss. In addition, it is very suitable for millimeter-wave and terahertz interconnections [9]. Therefore, the multi-layer filtering balun will have lower insertion loss and can be further interconnected and integrated with other active devices to obtain threedimensional integration [10]. This paper is organized as follows. Section II demonstrates the realization of the proposed multi-mode resonator with quarter-mode FSIW cavity. Section III presents an example of the filtering balun design. Section IV describes the fabrication process. Section V illustrates the measurement results and tolerance analysis. Section VI concludes the study.

II. Mode Analysis Of The Quarter-Mode FSIW
Two types of the quarter-mode FSIW can be obtained by edge and diagonal folding, as shown in Figs. 1 (a) and (b), respectively. Figs. 1 (c) and (d) show the top view of the quarter-mode edge and diagonal FSIW, respectively. For a quarter-mode edge FSIW cavity, a and b are the side lengths. In the following design, we set a = b. The boundary conditions of a quarter-mode edge FSIW cavity and their solutions of TMmn0 modes were presented in [5]. The eigenmode frequency in Fig. 1 satisfies: The modes in the following context are unified into a full-mode SIW cavity with a dimension of 2a × 2b.
The electric field distributions of the first five modes for the quarter-mode edge FSIW cavity are shown in Figs. 2 (a)-(e).   TE130 and TE310 are degenerated modes where the quality factor is higher than TE110. We modified the quartermode edge FSIW cavity to control the degenerated modes. Fig. 3 (a) shows the structure wherein a grounded metal wall is introduced in the open-circuit vertex of the quartermode edge FSIW resonator where a, b, and s are 2 mm,2 mm and 0.1 mm, respectively. E/2 is the distance between the grounded metal wall and the center of the quarter-mode edge FSIW cavity. Fig. 4 shows the relationship between normalized frequency and E/2 for the different modes of the quarter-mode edge FSIW resonator when only a grounded metal wall is introduced. Revised f4 (TE330) is the ratio between the resonance frequency (TE330) with metal wall at the open apex and the original resonance frequency (TE330) without metal wall at the open apex. It can be found that revised f4 (TE330) is decreasing when E/2 is increasing. The degenerate modes TE130 and TE310 can be separated if we decrease E. When E ≤ 2 mm, the grounded metal wall will not much affect the degenerated modes where the curves become flat. Therefore, E = 2 mm can be used as an initial value for design optimization. Table 1 presents the normalized frequency for the modified quarter-mode edge FSIW.
The frequency gap between TE130 and TE330 mode is approximately 1.2 times when E = 2 mm. A slot line is loaded to tune the TE330 mode. It is at the midpoint of the metal wall and is parallel to the edge of the quarter-mode edge FSIW, as shown in Fig. 3 (b). Fig. 5 shows the electric field distributions of TE130 and TE330 after loading the slot line. Fig. 6 shows the normalized frequency of each mode for the quarter-mode edge FSIW versus L/2 where E is set to 2 mm. TE330 mode move toward TE130 and TE310 by increasing parameter L. When L is 1.94 mm, the frequencies of TE130 and TE330 are getting close. The frequencies of TE310 and TE130, as well as the fundamental mode of TE110, increase when L is smaller than 1.94 mm. On the contrary, they decrease if L is larger than 1.94 mm. The fundamental mode TE110 is about 1.5 and 1.9 times lower than TE310 and TE130 when L is 1.94 mm because TE110 increases slower than TE310 and TE130. Therefore, quarter-mode edge folded FSIW has good suppression if we use TE130, and TE330 to design a balun. The electric field distributions of the quarter-mode diagonally FSIW cavity can also be plotted. Fig. 7 shows the first three modes. The TE220 mode provides the possibility for balun's design because the electric fields on both sides are equal in intensity and opposite in direction. If we set the appropriate dimensions of a and b in Fig. 1 (d), then the frequency of TE220 can operate at 90.43 GHz, whereas TE110 and TE130 are at 44.86 GHz and 100.91 GHz, respectively. The resonant frequency of TE130 is 1.17 times of TE220. Therefore, suppression should be considered. A grounded via is placed at the vertex of the quartermode diagonally FSIW to move TE130 mode away from TE220 mode. Fig. 8 shows the electric field distributions of the modified quarter-mode diagonally FSIW with a grounded via. Table 2 presents the comparisons of the normalized frequency. TE220 has less influence by setting a ground via while the normalized resonant frequency of TE130 has increased from 1.17 times to 1.31 times. This article has been accepted for publication in IEEE Access. This is the author's version which has not been fully edited and content may change prior to final publication. Citation information: DOI 10.1109/ACCESS.2022.3183579

III. Filtering Balun Design Example Using Quarter-mode FSIW
A W-band filtering balun with a center frequency f0 at 92 GHz and a fractional bandwidth (FBW) of 3.3 % is designed using the proposed quarter-mode FSIW. Fig. 9 (a) shows the structure of the filtering balun. It consists of a balun and two multimode resonators. Fig. 9 (b) shows the topology with coupled balance ports. Particle swarm optimization algorithm [11] is used to synthesize the coupling matrix of a third-order filtering balun. According to the symmetry of filtering balun topology, it can be concluded that: 12 14 13 15 22 44 (2) where the positive and negative signs of the coupling coefficient are according to the direction of the electric fields.
Through the above relationship, the degree of freedom of the adjacency matrix can be reduced, which reduces the convergence time of the algorithm to a certain extent. We improve the objective function [12] to form an objective function suitable for the extraction of coupling matrix of filtering balun: (3) Where   zi and   pi represent the zeros and poles of the filter function, respectively. The S parameters of filtering balun corresponding to symmetrical topology should meet the following requirements: (4) Therefore, Equation (3) can be simplified to: The coupling matrix is shown in Fig. 10 where a thirdorder filter response is designed with a transmission zero at 89 GHz, the in-band return loss is less than -20 dB. Fig. 11 shows the idea response of the proposed filtering balun, where the coupling matrix synthesis results show agreements with the simulated results and M12 and M13 are opposite according to the direction of the electric fields..  The structure of the proposed filtering balun is mainly composed of a modified quarter-mode diagonally FSIW cavity (a balun) and two modified quarter-mode edge FSIW cavities (multimode resonators), as shown in Fig. 12. The quarter-mode diagonally FSIW cavity operates at TE220 as resonator 1, as shown in Fig. 9. The modified quarter-mode edge FSIW cavities operate at TE130 and TE330 as resonators 2, 3, and 4, 5 as shown in Fig. 9. The coupling coefficient of the cavities can be calculated from Equation (6).
Where f0i and f0j represent the resonance frequency of different resonators. When f0i = f0j, we will have Striplines are used as the coupling structures between the modified quarter-mode diagonally FSIW cavity and two modified quarter-mode edge FSIW cavities. The maximum coupling coefficient can be obtained as the electric field distributions of the modified quarter-mode diagonally FSIW cavity (a balun) and is strongest at the midpoint of its left and right-angle edges, as shown in Fig. 8 (a). Fig. 13 shows the coupling coefficient versus parameter t between the edge and diagonally FSIW cavities. The initial value of t is set to 0.1 mm based on the coupling coefficient of the matrix.  The two modified quarter-mode edge FSIW cavities are coupled through their metal walls, which are equivalent to the electric wall. The coupling coefficient can be modified by varying the distance (E) between the cavities. Fig. 14 shows the coupling coefficient versus the distance (E) when a and b are set to 2 mm. The coupling coefficient between the cavities decreases gradually when the distance (E) increases. Based on the coupling coefficient of the matrix, the initial distance (E) is set to 2 mm at 92 GHz and then optimized by using full-wave simulation to obtain its final value. Fig. 15 (a) show the external quality factor (Qe1) versus parameters (S1), and (F) at the input of the filtering balun. Qe1 decreases with the increase in S1 and F. However, the decreasing rate of Qe1 becomes smaller if S1 increases, indicating that the suitable value of Qe can be obtained by selecting the appropriate parameters of S1 and F from the coupling matrix. Fig. 15 (b) shows the external quality factor Qe2 and Qe3 versus parameters (S2) at the output of the filtering balun. Qe can be calculated by equations (8) and (9), where Qe1 is 22.19, and S1 and F are set to 0.65 mm at 92 GHz as the initial value before optimization, respectively. Qe2 and Qe3 are 30.18 and 75.3, respectively. S2 is set to 0.47 mm.
Based on the topology and structure of the filtering balun, a transmission zero is generated on the left side of the passband, which improves frequency selectivity. A semicylinder is designed on the side wall of the modified quarter-mode edge FSIW to control the position of the transmission zero flexibly. The transmission zero is controlled by the radius (r) of the semi-cylinder (Fig. 12). Fig. 16 shows the S parameter of the filtering balun under weak external coupling excitation. The transmission zero can be controlled by varying radius r. Fig. 17 shows the normalized resonance frequency of each mode versus the radius (r) of the modified quartermode edge FSIW, where rd is set to 0.77 mm. The semicircle grounded via has less influence for the frequency of the cavities, except the transmission zero of the filtering balun.  Fig. 18 (a) shows the multilayer structures of the proposed multilayer filtering balun, which is fabricated on a silicon wafer. With MEMS technology, BCB polymer (CYCLOTENE 4026-46) is utilized as dielectrics. It has a dielectric constant of 2.65 and a low dissipation factor of 0.0008. The permeability is 1. Table  3 listed the design parameters.  Two layers of dielectric, each with a thickness of 70 μm are fabricated to enhance the quality factor of the filtering balun. In our process, cavities are etched in the silicon substrate, which is the half of the filtering balun. The thickness of one-layer BCB is deposited in the cavities. This method reduces the degree of surface warping and improves the fabrication yields. In addition, because of many regular-arranged vias in this SIW based filtering balun, a new fabrication method of silicon-based pillars with metallized surface, which help to reduce the fabrication difficulties and improve its accuracy, are investigated. Fig. 19 illustrates the fabrication process. A 50 Ω thin film resistor [13] is also fabricated in the multilayer packaged filtering balun because of difficulties in three port millimeter wave measurements. Therefore, two filtering baluns are fabricated, one 50 Ω thin film resistor is terminated in the third ports when S21 is measured, by the contrast, one 50 Ω thin film resistor is terminated in the second ports, when S21 is measured.

IV. Fabrication of the Filtering balun
1) A 1000 μm thickness high resistivity silicon wafer is used as a substrate, as shown in Fig. 19 (a). 2) Vertical cavities and silicon-based pillars are etched using BOSCH process with photoresist protection, as shown in Fig. 19 (b). 3) A thin-film resistor (50 Ω) is fabricated by using sputtering, photolithography, and etching [13], as shown in Figs. 19 (c)-(e). 4) A layer of negative photoresist (5300) is spread, and photolithographic development is carried out for the graphic of metal layers, as shown in Fig. 19 (f). 5) A seed layer is then sputtered, and 4 μm copper is then electroplated, as shown in Fig. 19 (g).
6) Silicon-based pillars and the transfer of the metal layer pattern are completed after the wafer soaked in acetone for several minutes, the copper is then peeled off along with the photoresist 5300, as shown in Fig. 19 (h). 7) BCB is then spin coated several times until the BCB in the cavities is higher than the surface, as shown in Fig. 19 (I). 8) Mechanical polishing is then used to make the surface flat where the BCB is the same level of the silicon surface, as shown in Fig. 19 (j). 9) The electrode of the thin-film resistance is then exposed after a thin layer of BCB is spin coated and developed, as shown in Fig. 19 (k). 10) The second seed layer is then sputtered with patterns created, and metal layers are fabricated by electroplating and ion beam etching, as shown in Fig. 19 (l). 11) BCB is then spin coated several times to develop another 70 um thickness, as shown in Fig. 19 (m). 12) The third metal layer is then developed after sputtering, photolithography, electroplating, and ion beam etching, as shown in Fig. 19 (n).

V. Experimental Results and Discussion
The measured stage consists of a Keysight 5247A network analyzer, two Keysight N5250CX10 frequency extender modules, two Cascade GSG probes (I100-AM-GSG-100), a Cascade E300 probe station, and some accessories. This stage supports the measured frequency range from DC to 110 GHz. The photo of the measurement setup is shown in Fig. 20. Fig. 21 shows the simulated and measured S-parameter of the filtering balun where one of the output ports are terminated by a 50 Ω thin film resistor.
The measurement shifts approximately 1.1 GHz with the minimum insertion loss 3+1.8 dB, which is 0.9 dB higher than the simulated results. The out-of-band suppression of transmission zero is about 40 dB at 91 GHz, which is 1.18 GHz offset compared with the simulated results. Fig. 22 shows the simulated and measured amplitude and phase imbalance of the filtering balun. The measurement results are focused on their imbalance of 91.9-95.6 GHz because of the frequency deviation. The overall measured amplitude imbalance is less than 0.9 dB with the phase imbalance less than 10°. The measured amplitude and phase imbalance have differences of 0.4 dB and 3°compare to the simulated results. Table 4 shows the comparisons between our proposed filtering balun and other works. Our proposed work has advantages in insertion loss and size. The tolerance of the fabrication is important for the development of a real filtering balun. These factors often cause the frequency shift between the simulated and measured results. For example, the fabricated filtering balun in Fig. 19 has a center frequency shift of 1.1 GHz. The main reason for this shift is the fabrication inaccuracies. We mainly focus on the tolerance analysis of key dimension parameters where frequency shift might occur was carried out by HFSS in order to accurately predict the influence of different machining errors on the filter.  The electroplating thickness (dr) has impact to the frequency offset, but has less influence to the insertion loss and imbalance. For example, if the thickness increases to 10 μm, the frequency shifts about 0.37 GHz because the electroplating uniformly wraps a layer of copper on siliconbased pillars, and the diameter of the silicon-based pillars with metallized surface increases with the electroplating thickness. Therefore, the resonance cavity decreases when the resonant frequency increases.
The length (L) of the slot line will affect TM330 mode, if it is smaller than the designed value, then the frequency of TM330 mode will be higher, which will result in increase in the operating frequency of the filtering balun. In addition, if the width (Lk) of the slot line increases due to the increase in equivalent inductance, the center frequency of the filtering balun also shifts higher.

VI. Conclusions
In this paper, two types of modified quarter-mode FSIW have been analyzed and designed. A W band third-order multi-layer packaged filtering balun has been designed, simulated, and measured on the basis of these two types of FSIW. One layer 70 um thickness of BCB has been investigated with two layers has been achieved to obtain a low loss filtering balun based on our in-house MEMS fabrication process. The measured insertion loss of the filtering balun is 3+1.8 dB with a comparable small size. These types of packaged filtering balun are very suitable for the future 3D heterogeneous integration with other semiconductor devices.