Z-Shaped Metasurface-Based Wideband Circularly Polarized Fabry–Pérot Antenna for C-Band Satellite Technology

This research proposes a low-profile Z-shaped metasurface (MTS)-based wideband circularly polarized (CP) Fabry–Pérot antenna for C-band satellite communication. The proposed low-cost and low-complexity CP Fabry–Pérot antenna is realized by using three substrate layers: upper, middle, and lower. The substrates are of FR-4 type with a dielectric constant of 4.3 and loss tangent of 0.025. The upper substrate contains $9\times $ 9 periodically-arranged Z-shaped MTS unit cells functioning as the partially reflecting surface and circular polarization conversion, and at the center of the middle substrate sits a corners-truncated square patch. The lower substrate consists of a copper plate with an H-shaped slot at the center of the ground plane and a microstrip feed line. The lower and middle substrates function as the source antenna. The periodic Z-shaped MTS unit cells are utilized to enhance the impedance bandwidth (IBW) and gain of the source antenna and also to convert linearly polarized into CP wave. The antenna dimension is $1.5\lambda _{0} \times 1.5\lambda _{0} \times 0.51\lambda _{0}$ . Simulations are performed and experiments carried out. The measured IBW and axial ratio bandwidth are 64% (4.4– 7.6 GHz) and 18% (4.4– 5.3 GHz) at the center frequency of 5 GHz. In addition, the proposed antenna scheme achieves a measured 3-dB boresight gain bandwidth of 30% (4.3– 5.8 GHz) with the maximum gain of 12.88 dBic at 4.7 GHz, rendering the proposed Z-shaped MTS-based CP Fabry–Pérot antenna operationally suitable for satellite communication. In essence, the novelty of this research lies in the use of the low-cost and low-complexity Z-shaped MTS unit cell to effectively enhance the antenna gain and convert LP into CP wave.


I. INTRODUCTION
In satellite communication, circularly polarized (CP) patch antennas play an important role in receiving or transmitting signals in various planes. Unlike linearly polarized (LP) antennas, the CP patch antennas are less susceptible to fading or multipath interference. The CP patch antennas nonetheless suffer from low gain due to dielectric loss. To overcome the low gain inherent in the conventional CP patch antennas, several techniques have been proposed and incorporated into the CP antennas, such as CP antenna arrays [1], CP dielectric resonator antenna [2], and CP Fabry-Pérot antenna [3].
The associate editor coordinating the review of this manuscript and approving it for publication was Weiren Zhu .
Of particular interest are Fabry-Pérot antennas [4], [5] which could achieve higher gain with lower cost and less complexity, vis-à-vis the CP antenna arrays and CP dielectric resonator antenna. A typical Fabry-Pérot antenna consists of an LP source antenna (e.g., microstrip patch, dipole antenna), which is placed with air-gap cavity between the partially reflective surface (PRS) and the fully reflective ground plane.
The PRS is typically fashioned from periodically-arranged metasurface (MTS) elements functioning as the superstrate [6] to covert LP to CP wave [7] and to enhance the antenna gain [8]. The cavity between the PRS and the reflective ground plane affects the resonance characteristics of the antenna [9] and the antenna performance, including the impedance bandwidth (IBW), axial ratio bandwidth (ARBW), gain, and radiation efficiency.
As a result, this research proposes a compact Z-shaped MTS-based wideband CP Fabry-Pérot antenna for C-band satellite communication system. The proposed antenna is realized using three layers of substrates: upper, middle, and lower, with air gaps between substrates. The upper substrate contains 9 × 9 periodically-arranged Z-shaped MTS unit cells which function as the PRS to enhance the antenna gain and bandwidth. At the center of the middle substrate sits a corners-truncated square patch, and the lower substrate consists of the H-shaped slot ground plane and a microstrip feed line. The lower and middle substrates function as the source antenna. Simulations are performed to optimize the antenna parameters using CST Studio Suite, and an antenna prototype is fabricated and experiments carried out in an anechoic chamber. In this research, the performance metrics of the proposed antenna include the IBW, ARBW, gain, radiation pattern, and radiation efficiency.

II. ANTENNA DESIGN PROCESS
A. Z-SHAPED MTS ELEMENT DESIGN Figure 1 illustrates the schematic view of the Z-shaped MTS-based CP Fabry-Pérot antenna. The proposed antenna consists primarily of the Z-shaped MTS (functioning as the partially reflective surface), square patch (functioning as the radiating patch), and H-shaped slot ground plane. The radiating patch on the middle substrate radiates electromagnetic (EM) waves to the partially reflective surface (PRS) on the upper substrate, while some of the EM waves are reflected back by the ground plane (on the lower substrate) into the cavity.
The height of the cavity (h 0 ) is approximated at half wavelength at the center frequency (5 GHz) [18], [19], which is approximately 33 mm. To derive the cavity height with wide IBW and high gain, optimization was carried out for the optimal cavity height whereby h 0 was varied between 33, 34, and 35 mm. The optimal h 0 is 34 mm, which corresponds to the surface wave resonance.
In addition, to achieve wide 3-dB boresight gain bandwidth, the reflection phase of PRS must be positive. The directivity of the MTS-based Fabry-Pérot antenna is enhanced by PRS whose directivity can be calculated by using equation (1) [20].
where D is the directivity of the antenna and R is the magnitude of reflection coefficient of PRS. VOLUME 10, 2022 The evolution of the Z-shaped MTS unit cell involves two stages: the MTS unit cell of square shape and the MTS unit cell of Z-character shape, as shown in Figure 2. The upper substrate to which the Z-shaped MTS unit cells are affixed is realized by stacking one substrate (upper-stack substrate) on top of the other substrate (lower-stack substrate) without air gap. Both upper-and lower-stack substrates are of FR-4 with 12 mm × 12 mm (p × p) in dimension (0.174λ 0 × 0.174λ 0 ) and 1.6 mm in thickness for h t1 and h t2 . The dielectric constant (ε r ) and loss tangent (tan δ) are 4.3 and 0.025. Figure 2(a) shows the square-shaped MTS unit cell on the upper substrate layer of the proposed antenna scheme, consisting of the square-shaped element (W u ) on the upper-stack substrate and the square-shaped ground plane (W l ) on the lower-stack substrate. Figure 2(b) shows the simulated model of the square-shaped MTS unit cell. The square-shaped ground plane on the lower-stack substrate receives the incident LP wave from the source antenna (port 1) and transmits to the square-shaped element on the upperstack substrate where the wave is radiated into free space (port 2).
However, the square-shaped MTS unit cell possesses low reflectivity and high transmittivity, rendering it less ideal for the PRS due to the resulting low gain. As a result, the squareshaped MTS unit cell is transformed into the Z-shaped MTS unit cell with high reflectivity and low transmittivity. Figure 2(c) shows the proposed Z-shaped MTS unit cell, consisting of the Z-shaped element and the square-shaped ground plane. The Z-shaped element has evolved from the square-shaped element, while the square-shaped ground plane and the upper-and lower-stack substrates remain unchanged. The function of the Z-shaped MTS unit cell is to convert LP to CP wave.
In the simulation, the boundary condition [21] of the Z-shaped MTS unit cell for x-and y-polarization is of periodic boundaries for xz-and yz-plane, while the Floquet ports (port 1 and port 2) are excited in the z-direction.
Figures 3(a)-(b) show the equivalent circuit models of x-and y-polarized electric fields, respectively [22]. The Z-shaped element, voids on either side of Z-shaped element, the coupling between the Z-shaped element and the squareshaped ground plane, and square-shaped ground plane serve as the inductor (L 1 ), series capacitor (C 1 and C 2 ), parallel capacitor (C 3 ), series inductor (L 2 ) and series capacitor (C 4 ) of the x-and y-polarized electric fields.
Figures 4(a)-(b) show the magnitude of the simulated reflection coefficient (S 11 ) and transmission coefficient (S 21 ) under variable width of the square-shaped element and the square-shaped ground plane (W u = W l ): 10, 11, and 12 mm. In Figure 4(a), with W u = W l = 10 and 11 mm, the magnitudes of S 11 are 0.14 and 0.1 at 5 GHz and 5.77 GHz, respectively. With W u = W l = 12 mm, the magnitude of S 11 approaches 1. In Figure 4(b), with W u = W l = 10 and 11 mm, the magnitudes of S 21 are 0.86 and 0.88 at 5 GHz and 5.77 GHz, respectively. With W u = W l = 12 mm,  the magnitude of S 21 is close to 0. With W u = W l = 11, the magnitude of S 11 of the square-shaped element with the square-shaped ground plane is the lowest around the center frequency of 5 GHz, and the corresponding magnitude of S 21 peaks in the vicinity of 5 GHz. The optimal W l is thus 11 mm. However, the square-shaped MTS unit cell possesses low reflectivity and high transmittivity, rendering it less ideal for the PRS due to the resulting low gain.
Figures 5(a)-(b) show the magnitude of the simulated S 11 and S 21 under variable width of the square-shaped element (W u ): 9, 10, and 11 mm. As W u increases, the magnitudes of S 11 and S 21 shift to low frequency. With W u = 10 mm, the magnitudes of S 11 and S 21 are 0.56 and 0.48 at 5 GHz, achieving high reflectivity and low transmittivity. The optimal W u is thus 10 mm. To realize the LP-to-CP conversion for the Z-shaped MTS unit cell, the magnitude of S 21 of x-and y-polarization waves at the center frequency of 5 GHz must be identical (T x = T y ), and the corresponding phase difference between the magnitude of S 21 of x-and y-polarization waves (φ Tx − φ Ty ) is equal to ±90 • . The CP wave is characterized by the axial ratio (AR) which can be calculated by equation (2), as shown at the bottom of the page, [23]. where T x and T y are the transmission coefficients of the x-and y-polarization waves, respectively, and φ is the phase difference between the xand y-polarization waves.  Figure 2). In Figure 6(a), the magnitudes of S 21 associated with the xand y-polarization (x-and y-pol) waves for b = 2.5, 3.5, and 4.5 mm are identical: 0.57, 0.48, and 0.43 at 5.2, 5.06, and 4.7 GHz, respectively. In Figure 6(b), as b increased (2.5, 3.5, and 4.5 mm), the phase difference of the magnitude of S 21 between x-and y-pol waves at 5 GHz are 44 • , 88 • , and 121 • , respectively. Given b = 3.5 mm, the phase difference of the magnitude of S 21 between x-and y-pol waves at 5 GHz is 88 • . The optimal b is 3.5 mm.   Figure 2). In Figure 7(a), as c increases, the magnitudes of S 21 associated with the x-and y-pol waves for c = 5, 6, and 7 mm are identical: 0.52, 0.49, and 0.46 at 5.15, 5, and 4.85 GHz, respectively. In Figure 7(b), as c increased (5, 6, and 7 mm), the phase difference of the magnitude of S 21 between x-and y-pol waves at 5 GHz are 59 • , 89 • , and 119 • , respectively. Specifically, with c = 6 mm, the phase difference of the magnitude of S 21 between x-and y-pol waves at 5 GHz is 89 • . The optimal c is 6 mm. Specifically, the vertical length (a), the length of the MTS element with voids (p), and the square-shaped ground plane (W l ) of one unit of Z-shaped MTS unit cell are 10, 12, and 11 mm, respectively.
The polarization conversion ratio (PCR) of the proposed Z-shaped MTS unit cell is determined by the reflection coefficients of the co-and cross-polarization waves, as expressed in equation (3) [24].
(2) VOLUME 10, 2022 where R RHCP and R LHCP are the reflection coefficients of the co-and cross-polarization waves, respectively. Figure 8(a) shows the reflection coefficients of the co-and cross-polarization waves and the PCR of the Z-shaped MTS unit cell. The simulated reflection coefficient of co-polarization is below −10 dB between 3.6 -6 GHz, while the simulated reflection coefficient of cross-polarization is above −5 dB between 4.1 -5.9 GHz. The PCR is greater than 0.9 between 4.3 -5.2 GHz.
where C + and C − are the circular conversion coefficients for left-hand circular polarization (LHCP) and right-hand circular polarization (RHCP), respectively. T x and T y are the transmission coefficients of E i x and E i y [27]. In Figure 8  x-axis relative to the frequency, which can be calculated by equation (7) [28].
In Figure  The upper substrate contains 9 × 9 periodic Z-shaped MTS unit cells, and at the center of the middle substrate sits a corners-truncated square patch functioning as the radiating patch. The dimensions of the square patch and truncated corners are 20 mm × 20 mm (W p × L p ) and 7.5 mm (L c ). Meanwhile, the lower substrate consists of a microstrip feed line functioning as the signal input and the H-shaped slot at the center of the ground plane (i.e., the aperture-coupled microstrip patch antenna). A conventional rectangular-shaped slot is normally utilized to manipulate the input impedance to achieve the maximum coupling between the radiating patch and ground plane, giving rise to wide IBW [29]. Likewise, the input impedance could be manipulated by varying the slot shape, size, and position. In [30], an H-shaped slot is used to improve the coupling between the radiating patch and ground plane. Meanwhile, the effects of H-shaped slot parameters on the antenna performance are detailed in [31]. The lower and middle substrates function as the source antenna.
In Figure 12(a), given 7 × 7 and 9 × 9 unit cells, the IBW are almost identical and fall between 4.44 -7.48 GHz. With 11 × 11 unit cells, the IBW becomes slightly wider, covering 4.44 -7.7 GHz. In Figure 12(b), as the number of the Z-shaped MTS unit cells increases, the maximum gain also increases. The ARBW (AR ≤ 3 dB) of 7 × 7 and 9 × 9 unit cells are 6%, covering 4.5 -4.8 GHz; and 17.4%, covering 4.4 -5.27 GHz, as shown in Figure 12(c). However, with 11×11 unit cells, the ARBW is greater than 3 dB (AR > 3 dB) at 5 GHz. The optimal number of the Z-shaped MTS unit cells is thus 9 × 9 unit cells. Table 1 tabulates the parameters and optimal dimensions of the Z-shaped MTS-based CP Fabry-Pérot antenna.
The periodic Z-shaped MTS unit cells functioning as the PRS are utilized to enhance the bandwidth of the source antenna. The surface wave resonance [32] of the proposed Z-shaped MTS-based CP Fabry-Pérot antenna propagates on the PRS surface, resulting in high-order resonance mode. In Figure 13(a), the surface wave travels along the length of the PRS corresponding to the resonant length. The surface wave resonance can be calculated by equation (8) [33].
where β SW is the propagation constant of the surface wave resonance, L PRS , N , and p are the length of the PRS, number of the Z-shaped MTS unit cell, and length of the Z-shaped MTS unit cell. Figure 13(b) is the dispersion diagram of the Z-shaped MTS unit cell. The boundary conditions in the z-direction of the Z-shaped MTS unit cell are the perfect electrical conductor (PEC) in the xz and yz planes for Eigenmode Solver [34]. The resonance modes 1 -5 are simulated by CST under variable numbers of the Z-shaped MTS unit cells relative to the phase of surface wave.
With N = 7 (50.87 • ) and 9 (39.87 • ), the resonance mode 4 occurs at 6.7 and 6.2 GHz, respectively, improving the   resonance mode 3. However, with N = 11, ARBW at 5 GHz is greater than 3 dB due to the absence of resonance.
In Figure 14(a), the equivalent circuit diagram of the Z-shaped MTS unit cell [35] consists of an inductor (L M 1 ), series capacitors (C M 1 and C M 2 ), parallel capacitor (C M 3 ), series inductor (L M 2 ), and series capacitor (C M 4 ), representing a Z-shaped element, the voids between any pair of Z-shaped elements, the coupling between the Z-shaped element and the square-shaped ground plane, the squareshaped ground plane, and the voids between any pair of square-shaped ground plane elements, respectively.
In Figure 14(b), the equivalent circuit diagram of the corners-truncated square patch consists [36] of an inductor (L P ) and parallel capacitor (C P ), representing the square patch with truncated corners. Meanwhile, the equivalent circuit diagram of the microstrip feed line with the H-shaped slot ground plane [37] consists of an inductor (L F ), parallel capacitor (C F1 and C F2 ), inductor (L S1 ), parallel capacitor (C S1 ), parallel capacitor (C S2 ), parallel inductor (L S2 ), and resistance (R S ), representing the microstrip feed line and the H-shaped slot ground plane.
In Figure 14(c), the equivalent circuit diagram of the proposed Z-shaped MTS-based CP Fabry-Pérot antenna consists of the parallelly-connected equivalent circuits in Figures 14(a) and (b). The impedance of the Z-shaped MTS element with square-shaped ground plane (Z 1 ), the microstrip feed line (Z 2 ), H-shaped slot ground plane (Z 3 ), and cornerstruncated square patch (Z 4 ) can be calculated by equations (9) -(12), respectively.
Advanced Design System (ADS) simulation is used to optimize the circuit components (R, L, C) of the equivalent circuit of the proposed Z-shaped MTS-based CP Fabry-Pérot antenna. Table 2 tabulates the optimal circuit components of the Z-shaped MTS-based CP Fabry-Pérot antenna. Figure 15 compares the simulated IBW of CST and ADS simulation programs, and the simulation results are in good agreement. The lowest resonance frequencies (|S 11 | ≤ −10 dB) using CST simulation occur at 4.72 and 5.32, while those associated with ADS simulation are at 4.75 GHz and 5.43. CST simulation is used to optimize the antenna parameters that achieve very wide impedance bandwidth, while ADS simulation is utilized to verify the CST-generated optimal antenna parameters. Meanwhile, the equivalent circuit models are generated by ADS simulation and used to verify the CST simulation-generated optimal antenna parameters. Figure 15 shows the equivalent circuit model to characterize the impedance matching [38] of the proposed antenna scheme. Figures 15(a)-(b) compares the simulated resistance, reactance, and IBW of the proposed antenna scheme using CST and ADS programs. The CST and ADS simulation results are in good agreement. The lowest resonance frequencies (|S 11 | ≤ −10 dB) using CST simulation occur at 4.72 and 5.32 GHz, while those associated with ADS simulation are at 4.75 and 5.43 GHz. Figure 16 shows the surface current distribution with respect to phase variation (0 • , 90 • , 180 • , and 270 • ) on certain sections of the Z-shaped MTS-based CP Fabry-Pérot antenna at 5 GHz (the center frequency). The direction of polarization of the proposed Z-shaped MTS-based CP Fabry-Pérot antenna can be observed by the rotation of the electric field along the propagation direction. The electric field vectors travel in the +z direction and rotate clockwise, giving rise to right-hand circular polarization (RHCP).

III. EXPERIMENTAL RESULTS
A Z-shaped MTS-based CP Fabry-Pérot antenna prototype was fabricated and experiments carried out. Figures 17(a)-(b) show the front and rear of the Z-shaped MTS-based CP Fabry-Pérot antenna prototype for C-band satellite communication system. Due to the fabrication limitations, the four Z-shaped MTS unit cells at the four corners of the upper substrate are deliberately removed and replaced with four holes in order to securely mount the three substrate layers together. The bolts and nuts to mount the three substrate layers are of polytetrafluoroethylene (PTFE) Teflon spacer, which has no effect on the antenna performance. In the antenna assembly, the distances between substrate layers are measured by a digital vernier caliper Mitutoyo). Figure 18 shows the measurement setup in an anechoic chamber using a vector network analyzer (Rohde&Schwarz ZNLE6 model). A pair of ETS-Lindgren Model 3102 Series Conical Log Spiral antennas (i.e., the transmitting antenna)  are used to verify the right-hand (RHCP) and left-hand (LHCP) CP radiations of the prototype antenna (i.e., the receiving antenna). The far-field distance [39] between the transmitting and receiving antennas is 4 m. The experiments were carried out in two stages: first, calibrating the vector network analyzer using the calibration kit (ZV-Z235); and second, determining the performance of the prototype antenna.
The AR is determined by the co-polarization (co-pol) and cross-polarization (cross-pol) electric fields of the receiving antenna. The co-pol and cross-pol electric fields (E) VOLUME 10, 2022  between the transmitting and receiving antennas correspond to E co−pol and E cross−pol . In other words, the co-and cross-pol electric fields correspond to RHCP and LHCP radiations, respectively. As shown in Figure 14, the radiation pattern of the proposed Z-shaped MTS-based CP Fabry-Pérot antenna is of RHCP. The AR can be calculated by equation (13) [40] AR(dB) = 20 log where E co−pol and E cross−pol are the electric field magnitudes of the co-and cross-pol between the transmitting and receiving antennas. In Figure 19(a), the simulated and measured IBW at the first (4.2 GHz) and second resonance (4.7 GHz) are in good agreement. Nonetheless, the measured IBW at the third resonance slightly shifts to the lower frequency (from 5.3 GHz to 5.2 GHz). The shift is attributable to the removal of the Z-shaped MTS unit cells at the four corners of the upper substrate to make room for the Teflon spacers to mount the three layers of substrate. The fourth resonance occurs at 5.6 -5.75 GHz due to improved impedance matching. Essentially, the proposed Z-shaped MTS-based CP Fabry-Pérot antenna possesses very wide IBW, covering the frequency range of 4.4 -7.6 GHz, resulting in the discarding of the first resonance. Figures 20(a)-(c) show the simulated and measured RHCP and LHCP radiation patterns of the Z-shaped MTS-based CP Fabry-Pérot antenna in the xz-and yz-planes at 4.5, 5, and 5.2 GHz, respectively. The simulated and measured RHCP are in good agreement, while those of LHCP are satisfactorily agreeable. Figure 21 compares the simulated and measured 3-dB boresight gain bandwidth and radiation efficiency of the Z-shaped MTS-based CP Fabry-Pérot antenna. The simulated and measured 3-dB boresight gain bandwidths are 20% (4.4 -5.4 GHz) and 30% (4.3 -5.8 GHz), respectively. The simulated and measured maximum gain are 13.08 dBic at 4.7 GHz and 12.88 dBic at 4.7 GHz. The measured radiation efficiency is more than 50% over the operating frequency (4.4 -5.3 GHz) of the Z-shaped MTS-based CP Fabry-Pérot antenna. Besides, the enhanced measured 3-dB boresight gain bandwidths at 5.1 -5.25 GHz and 5. 6 -5.75 GHz are attributable to the higher radiation efficiency (4.6 -5.25 GHz) and to the improved impedance matching (5.6 -5.75 GHz), respectively. Table 3 compares the existing CP MTS-based Fabry-Pérot antennas and the proposed Z-shaped MTS-based wideband CP Fabry-Pérot antenna in terms of the IBW, 3-dB ARBW, maximum gain, feed technique, and electrical dimension.
In [11], the CP single-feed hexagonal-shaped patch using superstrate encounters narrow ARBW in spite of wide IBW and high gain. In [13], the Fabry-Pérot antenna with polarization conversion metasurface (PCM) achieves high gain but fails to significantly enhance IBW and ARBW. Furthermore, in [15], the high-gain CP dual-feed Fabry-Pérot antenna using PCM is restricted by very narrow IBW and ARBW. In [16], the Fabry-Pérot patch array antenna with Wilkinson power divider feed network using chessboard PCM suffers from narrow ARBW despite wide IBW and high gain. In addition, the Wilkinson power divider feed network requires resistors, making the feed network very complex. In [17], the 2 × 2 PCM slot antenna array using sequentiallyrotated feed network achieves wide ARBW and high gain but suffers from narrow IBW.
The wider IBW of the proposed antenna scheme, vis-à-vis those of the existing PRS-based antennas (Table 3), could be attributed to the periodically arranged Z-shaped MTS unit cells. The MTS unit cells function as the superstrate with high reflectivity and low transmittivity, resulting in the broadband IBW and high gain. Besides, the Z-shaped MTS unit cells convert LP to CP wave radiation with wide ARBW. Moreover, the H-shaped slot on the ground plane of the source antenna enhances the IBW. Meanwhile, the proposed low-profile Z-shaped MTS-based CP Fabry-Pérot antenna for C-band frequency spectrum could achieve wide IBW and ARBW as well as high gain, rendering the proposed antenna scheme operationally ideal for satellite communication system.

IV. CONCLUSION
This research proposes a Z-shaped MTS-based wideband CP Fabry-Pérot antenna for C-band satellite communication. The proposed antenna is realized using three layers of FR-4 substrate (upper, middle, and lower) with air gaps between substrates. The upper substrate contains 9 × 9 periodicallyarranged Z-shaped MTS unit cells, and at the center of the middle substrate sits the corners-truncated square patch. Meanwhile, the lower substrate consists of the H-shaped slot ground plane and the microstrip feed line. In the operation, the H-shaped slot ground plane with the microstrip feed line generates an LP wave which is subsequently radiated by the corners-truncated square patch. The Z-shaped MTS unit cells function as the partially reflective surface and LP-to-CP wave conversion, giving rise to high antenna gain and wide bandwidth.
The proposed Z-shaped MTS-based CP Fabry-Pérot antenna is initially modeled and simulated by using CST Studio Suite, and an antenna prototype was fabricated and experiments undertaken. The polarization conversion ratio (PCR) of the Z-shaped MTS unit cells is greater than 0.9, with a bandwidth of 23% (4.25 -5.4 GHz) and normalized ellipticity of 1 (4.4 -5.2 GHz) for RHCP. In addition, the CST-and ADS-simulated impedance matching (|S 11 | ≤ −10 dB) of the proposed antenna scheme are between 4.5 -7.5 GHz. The measured IBW, ARBW, and 3-dB boresight gain bandwidth are 64% (4.4 -7.6 GHz), 18% (4.4 -5.3 GHz), and 30% (4.3 -5.8 GHz). The measured maximum gain and radiation efficiency are 12.88 dBic at 4.7 GHz and over 50% in the frequency range of 4.4 -5.3 GHz. The simulated and measured RHCP radiation patterns are in good agreement.
Due to low cost and light weight, compact satellites are increasingly adopted for low earth orbit (LEO) C-band satellite communication [41]. The compact satellites also require small antennas that are of low profile, low cost, high gain, wide IBW, and ARBW for transmitting and receiving signals to the ground station. The proposed low-profile Z-shaped MTS-based wideband CP Fabry-Pérot antenna for C-band frequency spectrum could achieve wide IBW and ARBW as well as high gain, rendering the proposed antenna scheme operationally suitable for satellite communication system.