http://ieeexplore.ieee.org TOC Alert for Publication# 8 2013 May 23 61 5 C1 2362 146 61 5 C2 C2 138 $0.38 lambda_{0}$ . Experimental results show that the omnidirectional CP antenna has bandwidths of 41% (1.65–2.5 GHz) for ${hbox {axial ratio}} < 3$ dB and 45% (1.58–2.5 GHz) for ${{hbox {return loss}} > 10}$ dB. The gain variation in the omnidirectional plane is less than 1 dB for the frequency range from 1.65 to 2.5 GHz. Good agreement is obtained between simulated and measured results.]]> 61 5 2363 2370 1817 61 5 2371 2379 3238 $({rm ZrO}_{2})$, and operates within the medical radio (MedRadio) band. An omnidirectional radiation pattern was obtained from simulations of human models to eliminate specific orientation dependence. A realistic measurement of an oral cavity was also performed under the Institutional Review Board (IRB) protocol to evaluate its practical biomedical effects. A miniature antenna with a total area of less than 245 ${rm mm}^{2}$ was designed and implemented. The measured performance achieved an antenna gain of $-$6.78 dBi, and had an impedance bandwidth of approximately 60 MHz. Therefore, a compact high-gain antenna with a large bandwidth was achieved.]]> 61 5 2380 2387 1533 61 5 2388 2396 2067 61 5 2397 2403 1549 61 5 2404 2410 1163 61 5 2411 2417 1537 $(beta =0)$ with non-zero group velocity at the zeroth-order resonance. Due to the unique property of an infinite wavelength, the current distribution of the dipole antenna can be improved. Furthermore, the MNG-TL dipole is also useful to achieve a wide impedance bandwidth. A prototype of the proposed MNG-TL dipole is fabricated and measured. It can provide a wide effective bandwidth of about 1.55 GHz (1.85–3.40 GHz) with VSWR 2:1 (${rm S}_{11}< -10$ dB) impedance matching and stable omnidirectional radiation patterns in the azimuth plane.]]> 61 5 2418 2424 1271 61 5 2425 2431 2348 ${rm H}$-plane irises, which produce a negative permittivity and negative permeability, respectively. To design a balanced CRLH transmission line, the ${rm H}$ -plane irises are materialized by twisted shapes that present a minimized coupling with the shorted stub. The independent controls of the LH and RH regions above the cutoff frequency make it relatively easy to obtain a balanced condition due to the minimized coupling effects. Additionally, using the CRLH waveguide, a beam scanning leaky-wave antenna (LWA) is designed. The LWA achieves good leakage radiation and high gain performance through insertion of the slot, which is set as close as possible to the operating frequency band. The design results show that the LWA supports the backward to forward beam scanning property, including a broadside radiation pattern by changing the frequencies without a band gap. The antenna consisting of 20 unit-cells has a beam steering angle from $-{hbox {31}}^{circ}$ or $+{hbox {52}}^{circ}$ and a high gains performance of more than 11.5 dBi in a frequency band from 9.5 GHz to 12.4 GHz.]]> 61 5 2432 2440 1508 $^{2}$ -K for all instrumented member geometries. Modeling with a simple lumped-thermal-element representation is presented, which accurately matches observed temperatures. The heat and ventilation equipment and controls are demonstrated to be sufficient to keep modeled antenna gain reductions due to thermally-induced distortions ${<}0.4$ dB more than 90% of the time during heating season. Gain reductions during summer will be dominated by vertical stratification of air in the radome, which is shown to be associated with daylight hours and outside temperature ${>}15^{circ}$C.]]> 61 5 2441 2448 2728 ${rm TE}_{0{n}}$ modes ( ${n}=$ 1–6) at an oversized, open-ended circular waveguide (C76-waveguide, ${hbox{inner diameter}}={hbox{27.79 mm}}$, 70 GHz) radiating into free space has been investigated theoretically employing two scattering matrix codes (SMCs), the finite-difference time-domain code EMPIRE and the uniform geometrical theory of diffraction (UTD) as well as the first time experimentally. The measurements utilized mode converters for generation of pure ${rm TE}_{0{n}}$ modes and a wavenumber spectrometer for mode analysis in the oversized waveguide. The total power reflection computed by EMPIRE is 4.1 to 13.4 dB lower than calculated from free-space wave and waveguide mode impedances. In all cases, most of the reflected power is carried by the backward traveling ${rm TE}_{06}$ mode, which is the mode closest to cutoff. Experiments are in very good agreement with theoretical results.]]> 61 5 2449 2456 2880 61 5 2457 2465 1483 61 5 2466 2473 1820 ${- 50} $ dB sidelobes has been adopted in the spatial-angular transform.]]> 61 5 2474 2481 1838 61 5 2482 2489 1538 61 5 2490 2499 2681 61 5 2500 2510 1886 $({hbox{VSWR}}leq {hbox{2{:}1}})$. Previous empirical approximations for the minimum height of a wideband array and maximum bandwidth of a tightly coupled dipole array (TCDA) are compared with the theoretical limits and found to be in good agreement.]]> 61 5 2511 2518 1312 61 5 2519 2531 3283 $Omega$ load, 5.78 GHz operation frequency, and 25 mW receiving power. For the communication port, the measured reflection coefficient is $-$18 dB at 6.1 GHz center frequency, the gain is 7.0 dBi, and the cross polarization in the broadside direction is $-$ 15 dB.]]> 61 5 2532 2539 1691 ${approx}{hbox{6 kJy}}$ zenith system equivalent flux density), large instantaneous bandwidth (up to 78 MHz), and four independently steerable beams utilizing digital “true time delay” beamforming. This paper summarizes the design of LWA1 and its performance as determined in commissioning experiments. We describe the method currently in use for array calibration, and report on measurements of sensitivity and beamwidth.]]> 61 5 2540 2549 1467 61 5 2550 2560 1305 61 5 2561 2569 1997 61 5 2570 2588 5704 61 5 2589 2596 2303 61 5 2597 2604 1741 61 5 2605 2614 2244 61 5 2615 2622 2530 61 5 2623 2629 2051 61 5 2630 2638 1962 61 5 2639 2645 2023 $R$ , the distance $D$ between the two fibers in the same panel, the distance $L$ between the two fibers in double panel), the incident wave polarization angle and the fiber electric conductivity are investigated. respectively. Numerical results illustrate composite materials have frequency-selective behaviors and multilayer composite materials have a good shielding effectiveness in the lower frequency range.]]> 61 5 2646 2654 2212 61 5 2655 2663 1923 61 5 2664 2675 3113 61 5 2676 2687 3296 61 5 2688 2694 1358 61 5 2695 2707 4248 $J_{v}(z)=sum _{k}alpha _{k},S(beta _{k}{z})$ , where $alpha _{k}$ and $beta _{k}$ are constants and $S$ is a sinusoidal function. Using the new expansions, varying levels of accuracy and range of applicability can be achieved by varying the number of terms in the expansions. For example, a four term expansion of $J_{0}(z)$ yields an average relative error of $< .1%$ for $vert zvert leq {2pi }$ and same accuracy is achieved for an eight term expansion for an extended range $vert zvert leq {5pi }$. Further, a phase and amplitude corrected large argument asymptotic formula is studied such that, the lower limit of its usage is reduced to medium magnitude ranges of arguments. The new set of formulas can not only be incorporated into math libraries very easily but also be useful for treatment of radiation and scattering problems involving Bessel functions.]]> 61 5 2708 2713 1317 61 5 2714 2726 3019 , is designed to optimally handle layered cylinders where the number of layers is extremely large as might be encountered in structures designed using transformational optics. The method is also tractable for any stratification profile without the need for individual layer analysis. We implement a recently developed numerical method to calculate Bessel functions of complex order and argument. Our method is verified by comparison with previously published results. We also present new results for a 5-layer case demonstrating self consistency and improved accuracy over published methods. Finally, to illustrate the method's benefits, we present a brief analysis of two multilayer structures; a multilayer variation of the coated cylinder presented in and an example of a 7-layer case that approximates a material parameter gradient.]]> 61 5 2727 2734 1477 $O(sqrt{t})$.]]> 61 5 2735 2744 2297 $N$-dimensional search space following Newton's second law of motion, which is also used to describe the motion of air parcels within the earth's atmosphere. Compared to similar particle based algorithms, WDO employs additional terms in the velocity update equation (e.g., gravitation and Coriolis forces), providing robustness and extra degrees of freedom to fine tune. Along with the theory and terminology of WDO, a numerical study for tuning the WDO parameters is presented. WDO is further applied to three electromagnetics optimization problems, including the synthesis of a linear antenna array, a double-sided artificial magnetic conductor for WiFi applications, and an E-shaped microstrip patch antenna. These examples suggest that WDO can, in some cases, out-perform other well-known techniques such as Particle Swarm Optimization (PSO), Genetic Algorithm (GA) or Differential Evolution (DE) and that WDO is well-suited for problems with both discrete and continuous-valued parameters.]]> 61 5 2745 2757 2193 61 5 2758 2766 1469 ${rm return loss} > 15$ dB, covering the frequency range 1.65–2.85 GHz for 2G/3G/LTE applications. The antenna gain of the broadband antenna element is about 9 dBi. A ${pm}45^{circ}$ dual-polarized planar antenna consisting of two broadband antenna elements is proposed, which achieves a bandwidth of about 50% and an isolation of 30 dB. Two 8-element antenna arrays are developed respectively for the broadband antenna and for the ${pm}45^{circ}$ dual-polarized antenna. Both antenna arrays achieve a bandwidth of more than 58% (1.6–2.9 GHz). The antenna gains achieved for the both antenna arrays are higher than 15.5 dBi. The half-power beam widths in the horizontal plane for the antenna arrays are approximately $65pm 10^{circ}$, suitable for base station applications.]]> 61 5 2767 2774 2258 ${-}$23 dB at both bands. Antenna phase center variation remains less than 2 mm across both L1 and L2 bands. Furthermore, the performance comparison of the proposed CPRGP with the commercially available pinwheel antenna and the shallow corrugated ground plane is presented, showing the advantages of CPRGP for high precision GNSS applications.]]> 61 5 2775 2782 1155 61 5 2783 2791 1159 61 5 2792 2800 2504 61 5 2801 2806 1275 61 5 2807 2816 2392 61 5 2817 2828 3301 61 5 2829 2837 1585 $18times 19 {rm mm}^{2}$. The experimental results demonstrate that the proposed antenna is capable of operating over the 2.40–2.51 GHz, 3.35–3.94 GHz and 5.02–6.63 GHz frequency ranges, which is suitable for WLAN 2.4/5.2/5.8 GHz and WiMAX 3.5 GHz applications.]]> 61 5 2838 2841 917 $40times 15times 0.8 {rm mm}^{3}$, the proposed planar antenna has the compact operation with more than 30% antenna size reduction. Furthermore, the measured peak gains and antenna efficiencies are about 2.81/4.97 dBi and 79/87% for the LTE/GSM/UMTS bands, respectively.]]> 61 5 2841 2846 1162 ${rm SWR}<1.5$ from 1.55 to 2.87 GHz was achieved. Stable radiation pattern with 3 dB beamwidth 63.3 $pm$ 2.9 degree at H-plane and 63.4 $pm$ 2 degree at E-plane and an antenna gain of 9 $pm$ 0.6 dBi was found over the operating frequencies.]]> 61 5 2847 2852 1625 61 5 2852 2856 988 61 5 2857 2860 909 $phi$-direction provides translation in the radial direction, a radiating shape-shifting surface (RSSS) can be developed. To demonstrate the interesting characteristics of the approach, two resonant frequency-tunable antennas were prototyped. The antennas incorporate parasitic patches, which are repositioned over a fixed circular microstrip patch antenna. The mechanical movement results in tunability of the antenna resonant frequency without degrading the return loss bandwidth or radiation pattern. Experimental results for a design with electromagnetically-coupled parasitic patches demonstrate tuning of the resonant frequency by ${>}{10}%$ over the 2.7-3 GHz frequency range with essentially constant gain and return loss bandwidth. Greater than 26% tuning bandwidth is achieved with the second design in which the parasitic patches make electrical contact with the primary antenna element.]]> 61 5 2861 2864 918 61 5 2865 2867 596 61 5 2868 2873 2071 61 5 2874 2879 1630 61 5 2879 2883 697 61 5 2884 2884 87 61 5 C3 C3 104 61 5 C4 C4 330