<![CDATA[ IEEE Transactions on Antennas and Propagation - new TOC ]]>
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TOC Alert for Publication# 8 2017November 16<![CDATA[Table of contents]]>6511C15666226<![CDATA[IEEE Transactions on Antennas and Propagation]]>6511C2C296<![CDATA[A Low-Profile Wideband Substrate-Integrated Waveguide Cavity-Backed E-Shaped Patch Antenna for the Q-LINKPAN Applications]]>$2 times 2$ prototype is designed, fabricated, and measured for a demonstration. The measured results show that the prototype has a 10 dB impedance bandwidth of 34.4%, a gain of around 12.5 dBi with a narrow E-plane radiation beam within 37.5–46 GHz for long distance applications, and a gain of around 8 dBi with a broad E-plane radiation beam within 47–53 GHz for short distance applications. The proposed technique can be used to develop compact planar antenna for meeting both the short- and long-rang communication requirements of the emerging Q-LINKPAN wireless system.]]>6511566756765687<![CDATA[Dual-Band Magnetoelectric Dipole Antenna With Dual-Sense Circularly Polarized Character]]>$Gamma $ -shaped feeding structure. The shorted vertical plates work as quasi-magnetic dipoles (M-dipoles), while the connected horizontal plates work as electric dipoles (E-dipoles). Two M-dipoles and E-dipoles are designed to possess different heights and lengths, and thus they are able to achieve separate operation bands. To generate dual-sense CP performance, the M-dipoles and E-dipoles are placed in parallel and the adjacent E-dipoles extend along the opposite directions. Benefiting from the intrinsic 90° phase difference between the M-dipole and E-dipole, the left-hand CP is achieved within the lower operation band (2.15–3.4 GHz) and the right-hand CP is achieved within the upper band (4–6.3 GHz). The proposed antenna possesses a unidirectional radiation pattern with high front-to-back ratio ($ge 22$ dB). Furthermore, high gain ($ge 8$ dBi) and radiation efficiency ($ge 90$ %) are also achieved.]]>6511567756853259<![CDATA[On Methods to Determine Bounds on the $Q$ -Factor for a Given Directivity]]>$Q$ -factor for a given directivity for a small antenna of arbitrary shape. A higher directivity in a small antenna is closely connected with a narrow impedance bandwidth. The relation between bandwidth and a desired directivity is still not fully understood, not even for small antennas. Initial investigations in this direction have related the radius of a circumscribing sphere to the directivity, and bounds on the $Q$ -factor have also been derived for a partial directivity in a given direction. In this paper, we derive lower bounds on the $Q$ -factor for a total desired directivity for an arbitrarily shaped antenna in a given direction as a convex problem using semidefinite relaxation (SDR) techniques. We also show that the relaxed solution is also a solution of the original problem of determining the lower $Q$ -factor bound for a total desired directivity. SDR can also be used to relax a class of other interesting nonconvex constraints in antenna optimization, such as tuning, losses, and front-to-back ratio. We compare two different new methods to determine the lowest $Q$ -factor for arbitrary-shaped antennas for a given total directivity. We also compare our results with full electromagnetic simulations of a parasitic element antenna with high directivity.]]>6511568656961688<![CDATA[Compressed Dipoles Resonating at Higher Order Modes With Enhanced Directivity]]>6511569757011878<![CDATA[Gain-Enhanced Patch Antenna Without Enlarged Size via Loading of Slot and Shorting Pins]]>01 mode, this central line behaves as a virtual electric wall with zero electric field and maximum surface current underneath and on the patch, respectively. By doing so, the central slot serves as an extra radiator so that this slot-loaded patch produces radiation as an equivalent three-slot array. Meanwhile, a pair of shorting pins is longitudinally placed at the two symmetrical sides of this central slot to maintain the resonant frequency. Due to the out of phase between the electric fields in the central slot and two sided slots, the far-field radiated by the three slots can be canceled with each other in specific directions away from broadside. Thus, this cancellation results in sidelobe of low level in the E-plane due to emergence of radiation nulls, and further makes the main radiation beam more directionally selective. Accordingly, the directivity and gain of this proposed patch antenna are enhanced with no need of enlarging the overall size of patch. Finally, the simulated and measured results are provided to validate the operating principle and show that the directivity is increased by about 1.7 dB increment.]]>6511570257091828<![CDATA[Low-Profile Two-Arm Inverted-L Antenna Design for Vehicular HF Communications]]>$lambda /13 ~(text {L}) times lambda /30 ~(text {W}) times lambda /400$ (H) at the lowest frequency of operation. Antenna performance and achieved benefits under the condition of mechanical breakdown are also discussed. The scaled prototype of the proposed antenna mounted on a selected vehicular platform is built using additive manufacturing. Measurement results agree well with simulations thereby verifying the design procedure and all the discussed findings.]]>6511571057193729<![CDATA[A New Concept for a Flat Lens Design Using Dielectric Cylinders]]>6511572057312245<![CDATA[Dual-Port Planar Antenna for Implantable Inductively Coupled Sensors]]>in vivo measurement of physiological parameters in challenging locations of the human body. They are considered promising alternatives to the existing catheter-based biomedical transducers and battery powered implants. In this paper, we report a novel planar reader antenna for wireless interrogation of $LC$ -based passive implantable sensors. The antenna utilizes two planar, electromagnetically isolated loops for wireless detection of the resonance frequency of the $LC$ sensors. 2-D and 3-D versions of the antenna with the identical geometry were developed for a comparative study. A comprehensive analysis of the antenna together with in vitro evaluation of the antenna performance is presented. With the proposed antenna topology, the isolation level of 58 dB within the frequency range of 1–50 MHz, and at least 37 dB up to 100 MHz was achieved. The 3-D version of the antenna demonstrated improved performance by the T/R isolation of at least 64 and 44 dB within the frequency bands of 1–50 and 50–100 MHz, respectively.]]>6511573257392825<![CDATA[Generation of Circularly Polarized Conical Beam Pattern Using Torus Knot Antenna]]>6511574057463077<![CDATA[A Novel Boresight and Conical Pattern Reconfigurable Antenna With the Diversity of 360° Polarization Scanning]]>11 and TM_{02} resonant modes, which provides an innovative method of designing pattern reconfigurable antennas for the WLAN applications. The proposed method is validated by a good agreement between simulation and experimental results for an antenna designed to provide boresight and conical pattern switching at 2.4 GHz. The gain of 6.05 dBi and efficiency of 86.7% are measured in the boresight mode, corresponding to 4.39 dBi and 82.4% at conical mode.]]>6511574757563849<![CDATA[Optimized Manipulation of the Network Characteristic Modes for Wideband Small Antenna Matching]]>$N$ -port networks, which can be analyzed using the theory of network characteristic modes (NCMs). Due to the profound physical insights provided by an NCM, a desired antenna performance can be achieved by properly manipulating the modes through reactive loading at specified ports. This paper presents a new matching technique for an $N$ -port internally loaded small antenna. By combining the NCM with the differential evolution algorithm, optimal reactive load values can be calculated. These loads can manipulate the NCMs to match the antenna in a desired bandwidth. Unlike other matching techniques based on the NCM, the desired bandwidth is achieved through internal reactive loading without the need for an input matching network. Two examples of electrically small monopole antennas are studies. In both examples, we study the possibility to match the antennas in the GSM 900 band, and we further investigate the possibility to achieve wider bandwidths by varying the number of ports and the loading topologies based on the interpretation of the characteristic modes. The loaded antennas are wideband with a relatively stable radiation pattern and an efficiency higher than 90%. To confirm the theoretical results, a prototype is fabricated and measured.]]>6511575757674231<![CDATA[Wideband Triple- and Quad-Resonance Substrate Integrated Waveguide Cavity-Backed Slot Antennas With Shorting Vias]]>110 mode) is shifted upward and coupled with two higher modes (even and odd TE_{210} modes). As a result, a wide bandwidth with triple resonances is achieved for the antenna. Based on the similar principle, a quad-resonance antenna having an even wider bandwidth is also developed. Prototypes of the two antennas are fabricated and measured. With a low profile of $0.03lambda _{0}$ (wavelength in free space), the triple-resonance design has a bandwidth of 15.2% and a peak gain of 4.80 dBi, and the quad-resonance design has a bandwidth of 17.5% and a peak gain of 7.27 dBi.]]>6511576857754144<![CDATA[Fresnel Lens at Millimeter-Wave: Enhancement of Efficiency and Radiation Frequency Bandwidth]]>n-like radiation pattern. For this, an accurate technological process has been used to manufacture a lens that allows for smooth compensation of the phase shift. The lens is compared with a classic Fresnel lens having the same physical dimensions and feeding system. The improvement is validated by measurement, which has revealed a maximum measured gain of 38.9 dBi corresponding to a maximum measured aperture efficiency of 59% with a −2 dB radiation frequency bandwidth of 29.25 GHz around 90 GHz. This amounts to an aperture efficiency enhancement of 60% and −2 dB radiation frequency bandwidth enhancement of 72% as compared with those of the classic Fresnel lens.]]>6511577657862853<![CDATA[Analysis and Design of Tapered Uniaxial-Anisotropic Dielectric Rod Antennas]]>6511578757952007<![CDATA[A First-Order Model of the Multiple-Feed Toroidal Magneto-Dielectric Antenna]]>6511579658071761<![CDATA[Sparse Array Quiescent Beamformer Design Combining Adaptive and Deterministic Constraints]]>6511580858182334<![CDATA[Multilevel Subarray Modularization to Construct Hierarchical Beamforming Networks for Phased Array of Antennas With Low Complexity]]>6511581958281816<![CDATA[Perpendicular-Corporate Feed in Three-Layered Parallel-Plate Radiating-Slot Array]]>$16 times 16$ slot array antenna is designed for uniform excitation with the perpendicular and the planar corporate feeds, and fabricated in the 60 GHz band. At the design frequency of 61.5 GHz, the measured directivity is 33.5 dBi with the aperture efficiency of 90.6%.]]>6511582958362799<![CDATA[A 60 GHz Horizontally Polarized Magnetoelectric Dipole Antenna Array With 2-D Multibeam Endfire Radiation]]>$4 times 4$ SIW Butler matrix with a three-layered zigzag topology is then designed, which enables a size reduction of 45% for the matrix compared with conventional single-layered configuration but not affecting its operating characteristics. By employing a $2 times 4$ ME-dipole array with 90° twists, two-folded Butler matrices and four SIW 3 dB E-plane couplers, a multibeam endfire array that can radiate eight beams scanning in two dimensions is designed at the 60 GHz band. The fabricated prototype verifies that a wide impedance bandwidth of 22.1%, gain varying from 10 to 13 dBi and stable radiation beams can be obtained. Due to good performance and the compact structure with low fabrication costs, the proposed design would be attractive for future millimeter-wave wireless applications including 5G communications and the WiGig system.]]>6511583758453665<![CDATA[Scanning Enhanced Low-Profile Broadband Phased Array With Radiator-Sharing Approach and Defected Ground Structures]]>6511584658543845<![CDATA[94 GHz Substrate Integrated Waveguide Dual-Circular-Polarization Shared-Aperture Parallel-Plate Long-Slot Array Antenna With Low Sidelobe Level]]>$15 times 15$ shared-aperture parallel-plate long-slot array antenna. This multilayer topology has a smaller size compared with the single-layer design. A simple and feasible method is applied to control the radiation pattern, which is able to realize dual-CP low SLL beams without a complicated feeding network. Two 1-D sixteen-way unequal dividers are employed to suppress the SLL in two planes of a CP array. Then, a 90° coupler is employed in the feeding network to switch the polarization modes between left-hand circular polarization (LHCP) and right-hand circular polarization (RHCP). Finally, a prototype of SIW parallel-plate long-slot array antenna is fabricated. Simulation and measured results show that SLLs of the fabricated antenna are under −18.5 dB in two planes.]]>6511585558612841<![CDATA[3-D Coverage Beam-Scanning Antenna Using Feed Array and Active Frequency-Selective Surface]]>6511586258704019<![CDATA[Antenna Loss and Receiving Efficiency for Mutually Coupled Arrays]]>6511587158771884<![CDATA[Improved Broadband Matching Bound]]>6511587858851660<![CDATA[Ultrathin Broadband Absorber Using Frequency-Selective Surface and Frequency-Dispersive Magnetic Materials]]>$0.031lambda _{L}$ against the thickness of $0.114lambda _{L}$ of conventional FSS-based absorber.]]>6511588658942607<![CDATA[Implementation of Radiating Aperture Field Distribution Using Tensorial Metasurfaces]]>6511589559073198<![CDATA[3-D Absorptive Frequency Selective Reflector for Antenna Radar Cross Section Reduction]]>6511590859172605<![CDATA[Evolutionary Approach to Solve a Novel Time-Domain Cavity Problem]]>$nabla times $ and the boundary conditions over perfectly conducting cavity surface. The basis elements are derived with their physical dimensions, i.e., volt per meter and ampere per meter. The other factor is a scalar dimension-free time-dependent modal amplitude. A system of evolutionary equations (i.e., with time derivative) for the modal amplitudes is derived and solved explicitly under the initial conditions. The problem is solved in the Hilbert space of real-valued functions of coordinates and time.]]>6511591859311191<![CDATA[Bandwidth for the Equivalent Circuit Model in Square-Loop Frequency Selective Surfaces]]>6511593259392063<![CDATA[A 2-D FDTD Model for Analysis of Plane Wave Propagation Through the Reentry Plasma Sheath]]>$pi $ -shaped total-field/scattered-field (TF/SF) boundary to introduce the plane wave source to the simulation domain. Based on the current density convolution (JEC) method and the phase matching principle, six 1-D auxiliary propagators were generated to obtain field values at the TF/SF boundary. The convolutional perfectly matched layer was implemented as the absorbing boundary condition of the auxiliary propagator. The efficiency and accuracy of the proposed method were validated by numerical examples.]]>6511594059482419<![CDATA[Discrete Numerical Approach to the Fredholm Integral Method for Evaluating Scattering by Irregular Dielectric Particles]]>6511594959591686<![CDATA[A New Family of Exponential-Based High-Order DGTD Methods for Modeling 3-D Transient Multiscale Electromagnetic Problems]]>6511596059743887<![CDATA[FFT-Based Method With Near-Matrix Compression]]>6511597559832789<![CDATA[A Probabilistic Model for the Nonlinear Electromagnetic Inverse Scattering: TM Case]]>6511598459911348<![CDATA[Three-Site Diversity at Ka-Band Satellite Links in Norway: Gain, Fade Duration, and the Impact of Switching Schemes]]>6511599260013223<![CDATA[High-Resolution Microwave Breast Imaging Using a 3-D Inverse Scattering Algorithm With a Variable-Strength Spatial Prior Constraint]]>a priori information about the boundaries between different tissues in the breast into the inverse scattering algorithm. This spatial prior information can be derived from another imaging modality, such as magnetic resonance imaging. Our method exploits the fact that the dielectric properties within a tissue type exhibit low to moderate variability by favoring solutions to the inverse scattering problem, which have small variations in dielectric properties within each tissue region. The amount of variation tolerated in each regions is controlled by a spatial prior constraint parameter. We demonstrate the feasibility of the method by imaging detailed, anatomically inspired numerical 3-D breast phantoms. The performance in the presence of different levels of noise and for different choices of the constraint parameter is evaluated. We also demonstrate the robustness of the algorithm with respect to errors in the spatial prior information.]]>6511600260144013<![CDATA[Multipath Clustering and Cluster Tracking for Geometry-Based Stochastic Channel Modeling]]>6511601560283748<![CDATA[The Transition From Weak to Strong Diffuse Radar Bistatic Scattering From Rough Ocean Surface]]>$it R_{a}$ . Using them for the case of weak, or even moderate diffuse scattering that is characterized by $it R_{a} lesssim 1$ would lead to an incorrect result. At the same time, for practical applications, it is important to describe the transition from partially coherent scattering to completely noncoherent, strong diffuse scattering in terms of the bistatic radar cross section $sigma _{0}$ . This situation may occur for scattering of global navigation satellite system (GNSS) signals or other signals of opportunity from a rough ocean surface under weak winds, or from ice. In this paper, we obtain an expression for $sigma _{0}$ for the case of weak-to-moderate diffuse scattering using a formulation based on the small slope approximation of the first order. For a reasonable range of $it R_{a}$ , calculations can be quickly performed using standard desktop computers. We demonstrate results of such calculations for bistatic radar scattering for $L$ -band at low-to-moderate $it R_{a}$ and its transition to $sigma _{0}$ at large values of $it R_{a}$ .]]>6511602960341365<![CDATA[Electromagnetic Scattering From Individual Crumpled Graphene Flakes: A Characteristic Modes Approach]]>6511603560472776<![CDATA[Greene Approximation Wide-Angle Parabolic Equation for Radio Propagation]]>6511604860562540<![CDATA[Electrically “Long” Dipoles in a Collocated/Orthogonal Triad—for Direction Finding and Polarization Estimation]]>$L$ ) is under (1/10) of a wavelength $lambda $ . Such electrically “short” dipoles have very small input impedances, rendering such “short” dipoles to be inefficient radiators. Practical dipoles, with an electrical length of $({L}/{lambda }) in [{0.1, 1}]$ , have notably larger input impedance, hence making them better radiators. Of such practical dipoles, this paper investigates their use for azimuth-polar direction finding and for polarization estimation. This paper will first present the measurement model (i.e., array manifold) of a triad of such practical dipoles, collocated in space and orthogonally oriented. This paper will then develop the corresponding closed-form algorithms to estimate the bivariate azimuth–elevation direction-of-arrival or the bivariate polarization. Such closed-form algorithms previously have been unavailable in the existing literature for such a triad of electrically “long” dipoles of pragmatic radiation efficiency.]]>6511605760671962<![CDATA[Electrical Balance Duplexer Field Trials in High-Speed Rail Scenarios]]>balancing impedance must adaptively track time-domain antenna impedance variation. A rail scenario presents a potentially demanding use case for an EBD, as fast moving trains in the vicinity of the antenna may cause dynamically changing reflections, which can be received as self-interference. In this paper, measured dynamic antenna reflection coefficients at 745 and 1900 MHz from train mounted antennas are included in the EBD circuit simulations in order to investigate the resulting variation in Tx–Rx isolation, and determine requirements for balancing impedance adaptation. This paper also presents the results from rail-based field trials of a hardware prototype EBD, which implements real-time antenna impedance tracking. Results show that the rail scenario does result in variation in Tx–Rx isolation, but that rebalancing the EBD at the intervals of 5 ms was sufficient to maintain >50 dB isolation for ~95% of the time.]]>6511606860753293<![CDATA[Off-Body Antenna Wireless Performance Evaluation in a Residential Environment]]>6511607660841987<![CDATA[Analysis of Antenna Structure for Energy Beamforming in Wireless Power Transfer]]>6511608560942834<![CDATA[Study on Pattern Distortion and DOA Estimation Performance of Crossed-Loop/Monopole Antenna in HF Radar]]>6511609561063464<![CDATA[Robust Ultraminiature Capsule Antenna for Ingestible and Implantable Applications]]>$Omega $ . The antenna is synthesized and miniaturized using a hybrid analytical–numerical approach, and then optimized to conform to the inner surface of a 17 mm long biocompatible encapsulation (7 mm diameter). The substrate is 50 $mu text{m}$ thick. The capsule antenna is analyzed both in simplified and anatomically realistic heterogeneous phantoms. It remains matched at common implantation sites and through the whole gastrointestinal tract. Enhanced robustness allows using the antenna for a wide range of in-body applications. Computed reflection coefficients and radiation performance both show good agreement with measurements. The far field is characterized with the direct illumination technique using an analog fiber optic link. The realized gain (measured max. value −19.6 dBi) exceeds the counterparts by about 3 dBi. The proposed antenna contributes to the further development of a new generation of miniature in-body devices that involve complex and dense integration of sensors, logic, and power source.]]>6511610761193204<![CDATA[Beamwidth Properties of Endfire 1-D Leaky-Wave Antennas]]>651161206125988<![CDATA[A Fern Fractal Leaf Inspired Wideband Antipodal Vivaldi Antenna for Microwave Imaging System]]>6511612661291482<![CDATA[Analysis of Sleeve Dipole Antennas Fed by Ferrite-Loaded Coaxial Cables for a Scaled-Down Cross-Borehole Radar]]>651161306133797<![CDATA[An Optically Tunable Cavity-Backed Slot Antenna]]>6511613461391084<![CDATA[On the Covariance Matrix and Diversity Performance Evaluation of Compact Multiport Antenna Systems]]>651161406144915<![CDATA[Dual-Band Orthogonally Polarized Single-Layer Reflectarray Antenna]]>$X$ - and $K$ -bands. The reflectarray element consists of a circular patch surrounded by a microstrip ring. Two curved phase delay lines are attached to the patch and two curved phase delay lines are attached to the ring. The ring and patch are designed for the operation at $X$ - and $K$ -bands, respectively. Phase delay lines attached to the ring are spatially rotated by 90° with respect to the delay lines attached to the patch. This produces a linear polarization at $X$ -band, which is perpendicular to the linear polarization produced at $K$ -band. This also provides a high polarization isolation, which enables by adjusting the phase of the elements at each band independent of the phase at the other bands. At both operation bands, the required phase shifts are obtained by varying the lengths of the phase delay lines. The proposed element provides more than 450° linear phase range over the bands 8.7–10.7 GHz ($X$ -band) and 18–20.4 GHz ($K$ -band). Measurement results show the maximum gain of 27.7 dB at 9.7 GHz with 25% 1 dB gain bandwidth and the maximum gain of 31.8 dB at 19.2 GHz wit-
18% 1 dB gain bandwidth. By proper arrangement of the elements in the array, the cross polarization is reduced. The measured efficiency is 63% at 9.7 GHz and 42% at 19.2 GHz with a center-fed design for the reflectarray.]]>6511614561502063<![CDATA[A New Interval Pattern Analysis Method of Array Antennas Based on Taylor Expansion]]>6511615161561635<![CDATA[Shaped Beam Synthesis Based on Superposition Principle and Taylor Method]]>651161576160958<![CDATA[Single-Layered Circularly Polarized Substrate-Integrated Waveguide Horn Antenna Array]]>$0.12lambda _{mathrm {mathbf {0}}}$ at the center frequency (1.524 mm) for 24 GHz system applications. It comprises of an integrated phase controlling and power dividing structure, two waveguide antennas, and an antipodal linearly tapered slot antenna. With such a phase controlling and power dividing structure fully integrated inside the horn antenna, two orthogonal electric fields of the equal amplitude with 90° phase difference are achieved at the aperture plane of the horn antenna, thus, yielding an even effective circular polarization in a compact single-layered geometry. The measured results of the prototyped horn antenna exhibit a 5% bandwidth (23.7–24.9 GHz) with an axial ratio below 3 dB and a VSWR below 2. The gain of the antenna is around 8.5 dBi.]]>6511616161662062<![CDATA[Polarization-Adjustable Planar Array Antenna With SIW-Fed High-Order-Mode Microstrip Patch]]>$2times 2$ antenna array integrated with single-layer SIW feeding network is designed and tested to illustrate this technique. The results manifest that the planar integrated $2 times 2$ antenna array features adjustable polarization with considerable gain. Furthermore, a two-layer scheme of orthogonal SIW feeding is developed, which would provide a more flexible topology for the scale-expandable array antenna with adjustable polarizations.]]>6511616761721202<![CDATA[Absorptive Frequency-Selective Reflection and Transmission Structures]]>$0.12lambda _{0}$ at the lowest absorption frequency. The second one is an AFST with wide absorption bands at both sides of its transmission window. It exhibits 54% and 76% for the lower and upper absorption bands, respectively, with 1 dB insertion loss in the transmission window. A good agreement is achieved between simulation and measured results.]]>6511617361782900<![CDATA[A Novel Meander-Line Polarizer Modeling Procedure and Broadband Equivalent Circuit]]>6511617961843089<![CDATA[Rainfall Rate Prediction for Propagation Applications: Model Performance at Regional Level Over Ireland]]>6511618561891165<![CDATA[A New Efficient Thickness Profile Design Method for Streamlined Airborne Radomes]]>6511619061951107<![CDATA[Corrections to “High-Gain Filtering Patch Antenna without Extra Circuit”]]>Table I and two typos in Fig. 14 in the communication [1] as follows.]]>651161966196457<![CDATA[Corrections to “A New Efficient Unconditionally Stable Finite-Difference Time-Domain Solution of the Wave Equation”]]>[1], $r_{x}$ , $r_{y}$ , and $textbf {G}_{textbf {M}}$ were mistyped. The correct values are begin{align*} r_{x}=&frac {sqrt {2} Delta t}{sqrt {varepsilon mu } Delta x} sin {left ({frac { k_{x} Delta x}{2} }right )}\ r_{y}=&frac {sqrt {2} Delta t}{sqrt {varepsilon mu } Delta y} sin {left ({frac { k_{y} Delta y}{2} }right )}\ mathbf {G_{M}}=&begin{matrix} left [{ begin{matrix} big (1+r_{x}^{2}big )(xi ^{2}+1)-2xi &quad -(2r_{x} r_{y}) xi \ -(2r_{x} r_{y}) xi &quad big (1+r_{y}^{2}big )(xi ^{2}+1)-2xi end{matrix} }right ]. end{matrix} end{align*} ]]>651161976197119<![CDATA[Correction to “Closed-Form Jones Matrix of Dual-Polarized Inverted-Vee Dipole Antennas over Lossy Ground”]]>[1, p. 33], the authors would like to make the following correction.]]>651161986198103<![CDATA[Corrections to “Relativistic Aspects of Plane Wave Scattering by a Perfectly Conducting Half-Plane With Uniform Velocity Along an Arbitrary Direction”]]>[1], there was a typo in (13), which should read as a scalar equation for the $ {z}$ -component of $tilde {mathbf {E}}{tilde {mathbf {H}}}$ in the formbegin{equation*} tilde {E}_{ {z}} {tilde {H}_{ {z}}} = F(tilde {xi }^{i}) , tilde {E}_{ {z}}^{i} {tilde {H}_{ {z}}^{i}} + F(tilde {xi }^{r}) tilde {E}_{ {z}}^{r} {tilde {H}_{ {z}}^{r}} end{equation*} from which the remaining field components may be obtained. The full vector equation for the electric field is then given by begin{align*} tilde {mathbf {E}}=&F(tilde {xi }^{i}) tilde {mathbf {E}}^{i} + F(tilde {xi }^{r}) tilde {mathbf {E}}^{r} + frac {e^{i pi /4}}{sqrt { 2 pi tilde {k} tilde {rho } sin tilde {theta }_{0} }} e^{i tilde {k} (tilde {rho } sin tilde {theta }_{0} -tilde { {z}} cos tilde {theta }_{0})} \[3pt]× bigg { cos frac {tilde {phi }-tilde {phi }_{0}}{2} left [{ left ({ tilde {mathbf {E}}^{i}_{0} cdot hat {tilde { {z}}} }right) hat {tilde { {z}}} - tilde {mathbf {E}}^{i}_{0} }right ] +sin frac {tilde {phi }-tilde {phi }_{0}}{2} , hat {tilde { {z}}} times tilde {mathbf {E}}^{i}_{0} \[4pt]&+,cos frac {tilde {phi }+tilde {phi }_{0}}{2} left [{ left ({ tilde {mathbf {E}}^{r}_{0} cdot hat {tilde { {z}}} }right) hat {tilde { {z}}} - tilde {mathbf {E}}^{r}_{0} }right ] +sin frac {tilde {phi }+tilde {phi }_{0}}{2} , hat {tilde { {z}}} times tilde {mathbf {E}}^{r}_{0} bigg } end{align*} where $tilde {mathbf {E}}^{i,r}_{0}$ denotes the incident (reflected) electric field ampli-
ude at the edge of the half-plane in frame $tilde {S}$ . In addition, $Z tilde {B}_{h}$ in (24) should read as $tilde {B}_{h}$ , yielding begin{align*} tilde {bar {bar {F}}}_{re}=&dfrac {tilde {Q}}{c} \× ! begin{bmatrix}! 0 !&quad ! -!tilde {B}_{e} sin dfrac {tilde {phi }}{2} !&quad ! tilde {B}_{e} cos dfrac {tilde {phi }}{2} !&quad ! 0 \ tilde {B}_{e} sin dfrac {tilde {phi }}{2} !&quad ! 0 !&quad ! 0 !&quad ! -tilde {B}_{h} sin dfrac {tilde {phi }}{2} \ -tilde {B}_{e} cos dfrac {tilde {phi }}{2} !&quad ! 0 !&quad ! 0 !&quad ! tilde {B}_{h} cos dfrac {tilde {phi }}{2} \ 0 !&quad ! tilde {B}_{h} sin dfrac {tilde {phi }}{2} !&quad ! -tilde {B}_{h} cos dfrac {tilde {phi }}{2} !&quad ! 0 end{bmatrix}!. end{align*} None of the above-mentioned remarks affect the calculations and results presented in [1].]]>651161996199121<![CDATA[Fuel Your Imagination]]>6511620062001746<![CDATA[IEEE Transactions on Antennas and Propagation]]>6511C3C3136<![CDATA[Institutional Listings]]>6511C4C4465