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TOC Alert for Publication# 22 2015May 21<![CDATA[Table of Contents]]>635C1C476<![CDATA[IEEE Transactions on Microwave Theory and Techniques publication information]]>635C2C2141<![CDATA[Dispersion Equations of a Rectangular Tape Helix Slow-Wave Structure]]>635144514563709<![CDATA[Determination of Normalized Magnetic Eigenfields in Microwave Cavities]]>635145714673049<![CDATA[Radial Transmission-Line Approach for the Analysis of Ring Loaded Slots in Circular Waveguide]]>635146814742055<![CDATA[Application of Coherence Theory to Modeling of Blackbody Radiation at Close Range]]>635147514882577<![CDATA[Computational Feasibility Study of Contrast-Enhanced Thermoacoustic Imaging for Breast Cancer Detection Using Realistic Numerical Breast Phantoms]]>635148915012310<![CDATA[Efficient Statistical Simulation of Microwave Devices Via Stochastic Testing-Based Circuit Equivalents of Nonlinear Components]]>635150215111512<![CDATA[On the Design of Gyroelectric Resonators and Circulators Using a Magnetically Biased 2-D Electron Gas (2-DEG)]]>63515121517900<![CDATA[Nonlinear Behavioral Modeling Dependent on Load Reflection Coefficient Magnitude]]> , as a parameter. Although a rigorous approach requires a full 2-D load–pull model to cover the entire Smith chart, simulation and experimental evidence have shown that such a 1-D model—that retains only amplitude information of the load reflection coefficient—can give accuracy close to that of a full 2-D load–pull model. Consequently, neglecting the phase constitutes an approximation that provides large benefits without appearing to lead to a severe compromise in accuracy. Furthermore, compared with traditional load-independent models, the new -dependent models provide a major improvement in model accuracy. After a discussion of the model extraction methodology, examples are provided comparing traditional load–pull -parameter models with the model presented in this paper. The new model not only provides consistently good accuracy, but also has a much smaller model file size. Along with the examples that display the ability of the new modeling technique to predict fundamental frequency behavioral, a second harmonic example is also provided. The modeling approach is also validated using measurements results.]]>635151815293066<![CDATA[A Novel Load Mismatch Detection and Correction Technique for 3G/4G Load Insensitive Power Amplifier Application]]> silicon-on-insulator field-effect transistor, which were integrated with a InGaP/GaAs HBT PA monolithic microwave integrated circuit into a single module. A PA module was implemented using more advanced impedance detectors having eight-phase regions, which was measured with WCDMA R’99 and 10-MHz 16QAM long-term evolution signals centered at 1.95 GHz for verification of the proposed idea. When compared to a conventional PA, excellent adjacent channel leakage ratio improvements of 12.7 and 8.3 dB, respectively, were achieved under output voltage standing-wave ratio (VSWR) of 2.5:1 for both applications. Moreover, the idea was extended for efficiency enhancement in a linearity spec-compliant impedance region, and the PA showed power-added efficiency improvements of 1.6% and 0.5%, respectively, under output VSWR of 2.5:1 for both applications.]]>635153015434006<![CDATA[A General Digital Predistortion Architecture Using Constrained Feedback Bandwidth for Wideband Power Amplifiers]]>635154415552652<![CDATA[Concurrent Dual-Band Digital Predistortion With a Single Feedback Loop]]> , in conjunction with a 2-D carrier co-located memory polynomial model in PA forward modeling. The concurrent dual-band DPD models for each band can then be extracted successfully according to the forward modeling results. Experimental tests have been performed for a commercial Mini Circuits amplifier and a 120-W peak power GaN base-station PA. Different signal combinations have been tested for both balanced and unbalanced output power operations in the dual bands. The results validate that the proposed method is able to achieve linearization performances comparable to those of the conventional two parallel feedback loops based technique. Even when a nonideal feedback loop with nonflat frequency responses in the two bands is used, the new method performs well when the feedback loop is calibrated in advance. This single feedback loop based concurrent dual-band DPD architecture is a strong candidate for future broadband dual-band transmitting systems.]]>635155615683329<![CDATA[Tunable 1.25–2.1-GHz 4-Pole Bandpass Filter With Intrinsic Transmission Zero Tuning]]> at 200 MHz from the passband. These TZs can also be moved from the upper stopband to the lower stopband without any auxiliary tuning elements to significantly change the stopband suppression. The high rejection filter is useful in wireless systems and cognitive radios with large interferers.]]>635156915782529<![CDATA[Tunable 4-Pole Noncontiguous 0.7–2.1-GHz Bandpass Filters Based on Dual Zero-Value Couplings]]> and mil, two filters with two separate frequency bands (2- and 4-pole: Filter A) and one filter with contiguous frequency tuning (4-pole: Filter B). The 2-pole filter results in an insertion loss of 2.9–4.7 dB with a tuning range of 0.75–1.09 and 1.70–2.18 GHz. Filter A with a 4-pole response results in an insertion loss 3.7–7.9 dB and a -dB bandwidth of 68–87 MHz with a tuning ranges of 0.77–1.10 and 1.70–2.10 GHz. Filter B shows an insertion loss of 4.4–6.6 dB and a -dB bandwidth of 39–68 MHz with a tuning range of 0.70–1.44 GHz. The application areas are in wideband tunable filters requiring contiguous and noncontiguous frequency coverage with bandwidth tunability in separate bands.]]>635157915861958<![CDATA[Wideband Differential Bandpass Filters on Multimode Slotline Resonator With Intrinsic Common-Mode Rejection]]>635158715941659<![CDATA[A 1.4–2.3-GHz Tunable Diplexer Based on Reconfigurable Matching Networks]]>635159516021893<![CDATA[Quality Factor of the Waveguide Re-Entrant Turnstile Junction Circulator]]>63516031608913<![CDATA[Simple Broadband Quasi-Optical Spatial Multiplexer in Substrate Integrated Technology]]>635160916202570<![CDATA[A Broadband 4.5–15.5-GHz SiGe Power Amplifier With 25.5-dBm Peak Saturated Output Power and 28.7% Maximum PAE]]> , output 1-dB compression point , and power-added efficiency (PAE) in the range from 12.8 to 15.7 dB, 18.8 to 23.7 dBm, 16.7 to 19.5 dBm, and 11.4 to 31.9%, respectively. Its peak saturated output power and maximum PAE are both obtained at 8.5 GHz. Second, to increase the output power, a PA consisting of two parallel broadband cells with a power combination is presented. This PA operates in the 4.5–15.5-GHz frequency range with measured gain, , and PAE in the range from 11 to 16.6 dB, 21.3 to 25.5 dBm, 18.7 to 21.7 dBm, and 11.9 to 28.7%, respectively. It achieves its peak saturated output power of 25.5 dBm at 8.5 GHz and its maximum PAE of 28.7% with an associated output power of 23.6 dBm at 6.5 GHz. Each of those two PAs achieves better performances than the state-of-the-art in broadband SiGe technology when comparing the output power level and efficiency.]]>635162116322618<![CDATA[Power Synthesis at 110-GHz Frequency Based on Discrete Sources]]>635163316442200<![CDATA[A 47.6–71.0-GHz 65-nm CMOS VCO Based on Magnetically Coupled <formula formulatype="inline"><tex Notation="TeX">$pi $</tex></formula>-Type LC Network]]> -type LC network in 65-nm CMOS is proposed. By configuring the switched negative-resistance unit, the VCO can oscillate at the even mode or the odd mode of the magnetically coupled -type LC network, thus the tuning range is widened without introducing the switch loss into the resonator. The proposed VCO achieves a measured continuous tuning range of 39%, from 47.6 to 71.0 GHz. The measured phase noise for a 47.6-GHz carrier at the even mode and 56.2-GHz carrier at the odd mode are and at 10-MHz offset with a corresponding of and , respectively. The measured phase noise in the whole frequency tuning range varies from to at 10-MHz offset, while the corresponding figure-of-merit (FOM) and vary from to and to 190.6 dB, respectively. The VCO core consumes 8.9–10.4-mA current from -
.0-V power supply and 320230 die area.]]>635164516572914<![CDATA[An FBAR/CMOS Frequency/Phase Discriminator and Phase Noise Reduction System]]> m CMOS technology integrating the discriminator and the phase noise reduction schemes, respectively. The 1.5-GHz discriminator shows phase noise floor of dBc/Hz at 20 kHz, dBc/Hz at 100 kHz, dBc/Hz at 1 MHz, and dBc/Hz at 4 MHz, while consuming 26 mW of power. The measured phase noise of the feedback reduction circuitry reaches the phase noise floor of the discriminator within the loop bandwidth, as expected.]]>635165816651744<![CDATA[Fully Integrated D-Band Direct Carrier Quadrature (I/Q) Modulator and Demodulator Circuits in InP DHBT Technology]]> m440 m including the RF and LO baluns.]]>635166616751859<![CDATA[Digitally Assisted CMOS RF Detectors With Self-Calibration for Variability Compensation]]> and a 33-dB dynamic range at 2 GHz after digital correction. The circuit occupies an area of and consumes a maximum of from a 1.2-V supply. The results are confirmed by measurements performed on ten samples.]]>635167616821536<![CDATA[A Hardware Efficient Implementation of a Digital Baseband Receiver for High-Capacity Millimeter-Wave Radios]]>635168316921835<![CDATA[Generalized Theory of the Thru-Reflect-Match Calibration Technique]]> ). The classical theory of the TRM calibration refers the calibration to the impedance of the load used as the match standard (), which is assumed as symmetrical. In practice, may be frequency dependent. Moreover, as the frequency increases, the symmetry condition of the load used as match standard may become difficult to preserve, thus affecting the reference impedance definition and reducing the accuracy of the calibration. In this paper, the -parameters matrix formalism is used to introduce for the first time a generalized TRM calibration theory, allowing the use of either symmetrical or nonsymmetrical loads of arbitrary impedance as the match standard.]]>635169316991444<![CDATA[Material Characterization of Arbitrarily Shaped Dielectrics Based on Reflected Pulse Characteristics]]>635170017091835<![CDATA[Millimeter-Wave Modulated-Signal and Error-Vector-Magnitude Measurement With Uncertainty]]> 0.5% for a 1-GSymbol/s 64-state quadrature-amplitude-modulated signal at 44 GHz.]]>635171017201616<![CDATA[Analysis, Blind Identification, and Correction of Frequency Response Mismatch in Two-Channel Time-Interleaved ADCs]]>635172117343371<![CDATA[A K-Band CMOS UWB Four-Channel Radar Front-End With Coherent Pulsed Oscillator Array]]> CMOS technology and the chip size is 3.9 mm by 1.7 mm. The output spectrum is centered on 26.0 GHz and satisfies the Federal Communications Commission spectral mask. The system attains the angular resolution of 9 in the range from to . Total power consumption is only , mainly used in the receiver and digital blocks.]]>635173517452229<![CDATA[Signal Detection and Noise Modeling of a 1-D Pulse-Based Ultra-Wideband Ranging System and Its Accuracy Assessment]]>635174617572378<![CDATA[Theoretical Energy-Conversion Efficiency for Energy-Harvesting Circuits Under Power-Optimized Waveform Excitation]]>63517581767481<![CDATA[A Multi-Band Stacked RF Energy Harvester With RF-to-DC Efficiency Up to 84%]]> -dBm input power homogeneously widespread over the four RF bands and reaches 84% at 5.8 dBm. The relative error between the measured dc output power with all four RF bands on and the ideal sum of each of the four RF bands power contribution is less than 3%. It is shown that the RF-to-dc conversion efficiency is more than doubled compared to that measured with a single RF source, thanks to the proposed rectifier architecture.]]>635176817782540<![CDATA[Expand your professional network with IEEE]]>635177917792210<![CDATA[Expand your network, get rewarded]]>635178017803525<![CDATA[IEEE Transactions on Microwave Theory and Techniques information for authors]]>635C3C3122